Matic Šerc BPSK MODULATED SATELLITE COMMUNICATION … · 2017. 11. 28. · Matic Šerc BPSK...

Fakulteta za elektrotehniko,računalništvo in informatiko

Smetanova ulica 172000 Maribor, Slovenija

Matic Šerc

BPSK MODULATED SATELLITE COMMUNICATION SYSTEM MODEL

Maribor, September 2013

BPSK MODULATED SATELLITE COMMUNICATION SYSTEM MODEL

Master Thesis

Student: Matic Šerc

Study programme: Telecommunications

Mentor: doc. dr. Iztok Kramberger

Co-mentor: Pedro P. Carballo

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THANKS TO...

I would like to thank to both mentors

who helped me with my Master Thesis

work, doc. dr. Iztok Kramberger and

Pedro P. Carballo, for mentorship and

support.

I would also like to thank to my parents

for support during my studies.

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Model satelitskega komunikacijskega sistema z BPSK modulacijo

Ključne besede: digitalno, faza, modulacija, satelit, komunikacija, model, pridobitev,

postaja

UDK: 621.39:004.77(043.2)

Povzetek

V magistrski nalogi je predlagan in modeliran komunikacijski sistem za Trisat misijo.

Sistem je predlagan na osnovi analize parametrov in zahtev dotične misije. Predlagana je

komunikacija od fizičnega do aplikacijskega nivoja med končnima računalnikoma v

vesoljski postaji in v zemeljski postaji, pri čemer smo se osredotočili predvsem na nižje

nivoje komunikacijskega protokolnega sklada, saj so le-ti znotraj danih okvirjev manj

standardizirani. Poglobljeno smo analizirali lastnosti fizičnega razširjanja signalov,

izgube v zvezi in potrebna ojačanja za parametre, ki so specifični za misijo Trisat. Razvili,

modelirali in prototipirali smo BPSK demodulator za zemeljsko postajo, izveden v FPGA

integriranem vezju. Poleg tega je v magistrski nalogi predlagana tudi zasnova zemeljske

postaje, ki omogoča povezavo več računalnikov s satelitom preko IP omrežja med

oddajno-sprejemno postajo in končnim računalnikom zemeljske postaje.

Trisat vesoljska postaja bo Cubesat satelit. Cubesat je mednarodni projekt, v katerem

sodeluje več kot 40 univerz, visokih šol in podjetij. Projekt se je začel leta 1999 kot plod

sodelovanja med Politehnično univerzo v Kaliforniji (Cal Poly) in Univerzo v Stanfordu,

saj sta univerzi med drugim razvili specifikacije in standarde za satelite, ki jih lahko

razvijejo univerze in v okviru Cubesat programa pošljejo v vesolje. Sateliti morajo med

drugim ustrezati tudi standardom glede dimenzij, in sicer je tipičen Cubesat velik

v

10x10x10 cm in težek do 1,33 kg, ali pa gre za 3U različico, ki je velika 10x10x30 cm in

težka največ 4 kg.

Trisat je vesoljska misija Univerze v Mariboru in njenih industrijskih partnerjev. Namen

misije Trisat je demonstracija vesoljske tehnologije in znanosti za izobraževalne,

znanstvene in tehnološke namene. Vesoljska postaja misije bo 3U Cubesat satelit, ki bo

fotografiral Zemljino površje v vidnem spektru, v spektru blizu infrardeče svetlobe, izvajal

spektralno analizo sončevega sevanja med vidno in infrardečo svetlobo, ter komuniciral z

zemeljsko postajo in avtonomno letel okoli Zemlje na višini med 600 in 1000 km.

Satelit bo z zemeljsko postajo komuniciral na radioamaterskih frekvencah, in sicer v treh

frekvenčnih pasovih: komunikacija na UHF (iz satelita proti Zemlji) in VHF (iz Zemlje

proti satelitu) pasu bo namenjena telekomandam in telemetriji za osnovno delovanje

satelita (gre za popolno dvosmerni, frekvenčno deljeni kanal) ne glede na usmerjenost

satelita, hitrejša komunikacija na S-band pasu pa bo namenjena hitrejšemu prenosu

informacij, ko bo satelit z usmerjeno anteno usmerjen proti Zemlji, s čimer bo zagotovljen

prenos podatkov za nalogo satelita. Točne frekvence v tem trenutku še niso potrjene,

izbrane frekvence pa so bile nekje med 144 in 146 MHz na VHF pasu, nekje med 435 in

438 MHz na UHF pasu in nekje med 2.4 in 2.45 GHz na S-band pasu. Na UHF in VHF

pasu bo pasovna širina komunikacij 15 kHz, na S-band pasu pa 50 MHz z uporabljeno

BPSK modulacijo.

Komunikacijski sistem sestoji iz sledečih komponent: računalnik na satelitu, ki skrbi za

aplikacijsko komunikacijsko opremo, oddajno-sprejemni sistem na satelitu, antene na

satelitu, pot ki jo bodo prepotovali signali skozi vesolje in atmosfero, oddajno-sprejemne

antene an Zemlji, pot za manipulacijo signala (ojačanje, ...) na Zemlji, oddajno-sprejemni

sistem zemeljske postaje, IP omrežje na Zemlji, ter kontrolni računalniki – strežniki na

Zemlji.

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PREDLAGANA ARHITEKTURA ZEMELJSKE POSTAJE

Predlagana arhitektura zemeljske postaje je sestavljena iz sledečih sklopov:

Poti za manipulacijo s signali, ki se zaključuje z anteno na eni strani in z

oddajno/sprejemno enoto na drugi strani. Na poti so signali ojačani, po potrebi preslikani

na nižje ali višje frekvence, ter prenešeni med anteno in oddajno/sprejemno enoto. Takšne

poti so štiri: ena za sprejem UHF, ena za oddajo VHF, ena za sprejem na S-bandu in ena

za oddajo na S-bandu.

Oddajno sprejemna enota, ki v smeri proti satelitu izvaja modulacijo in demodulacijo, ter

v primeru telekomand tudi protokol komunikacije s satelitom.

Enota je sestavljena iz FPGA vezja, ARM procesorja in ostalih komponent, kot je na

primer analogno digitalni pretvornik. Ostale komponente so povezane na FPGA, FPGA pa

je povezan z ARM procesorjem preko I2C, SPI in večnamenskih vodil. V FPGA je

predlagana izvedba demodulatorja, ki je opisana v poglavju izvedbe demodulatorja. ARM

procesor je zadolžen za izvajanje IP in aplikacijskih protokolov in komunikacijo s strežniki

Zemeljske postaje.

IP omrežje je uporabljeno za komunikacijo med oddajno-sprejemo enoto in Strežniki

Zemeljske postaje. Na ta način je mogoče z oddajno-sprejemno enoto komunicirati od

koderkoli v IP omrežju, kar je koristno za izobraževalne in demonstracijske namene.

Strežniki Zemeljske postaje poganjajo aplikacijsko opremo, preko katere je mogoče

pošiljati telekomande, nastavljati parametre komponent znotraj oddajno-sprejemne enote,

ter direktno pošiljati na satelit ali s satelita sprejemati podatke preko S-band satelitske

povezave. Za nastavljanje in branje parametrov oddajno-sprejemne enote je predlagana

uporaba SNMP protokola, za S-band komunikacijo se po IP omrežju predlaga uporaba

UDP protokola, za telekomande pa TCP protokola do oddajno-sprejemne enote, ki potem

telekomande enkapsulira v primeren format za pošiljanje na satelit.

Ker bo za pohitritev izvedbe zemeljske postaje za potrebe sprejema in oddaje signalov

uporabljen NI USRP-2920 univerzalni programsko definiran rajdijski sprejemnik in

oddajnik, ki deluje v frekvenčnem območju med 50 MHz in 2200 MHz, bodo signali iz S-

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band pasu preslikani na nižje frekvence, in sicer signali iz frekvenčnega območja med

2000 in 2700 MHz bodo preslikani na frekvence med 167 in 867 MHz na sprejemni veji.

Na oddajni veji bodo signali preslikani iz pasu med 597 MHz in 797 MHz na pas med

2300 MHz in 2500 MHz.

KOMUNIKACIJA NA RF NIVOJU

Satelitski komunikacijski sistem bo priključen direktno na antene na satelitu in potrebuje

vstopne nivoje signala med -120 in -70 dBm na VHF območju. Njegova izhodna moč na

UHF območju je med 27 in 33 dBm. Antene na satelitu na VHF in UHF območju ne bodo

imele dobitka (gain), zato bodo ojačanja signalov potrebna na Zemlji. Na S-bandu bodo

moči na izhodu med 21 in 30 dBm, vendar bo signal dodatno ojačan z usmerjeno anteno,

ki ima 6 dB dobitek (gain).

Signal bo potoval skozi vesolje in atmosfero med zemeljsko in vesoljsko postajo. Pri tem se

bo največ signala izgubilo na poti skozi prazen prostor, nekaj v ionosferi, nekaj v atmosferi

zaradi rotacije polarizacije, lomnih količnikov zemeljske atmosfere, Rayleighovega

presihanja in slabljenja zaradi vremena. Prišlo bo tudi do popačenj zaradi Dopplerjevega

efekta. V okolici zemeljske postaje bodo prisotni tudi drugi elektromagnetni signali,

dodaten šum pa se bo ustvarjal v anteni in komponentah v zemeljski postaji in bo odvisen

od temperature komponent.

Izračunane izgube na poti zaradi razdalje na S-band pasu 159,5 dB, na VHF pasu 135,8

dB in na UHF pasu 145,4 dB. Do dodatnih izgub bo prišlo zaradi nepopolne usmerjenosti

(„naciljanosti“) anten in zaradi atmosferskih izgub, tako da bo na frekvencah v S-band

frekvenčnem pasu nivo signala v okolici zemeljske postaje predvidoma okoli -130,2 dBm in

v UHF frekvenčnem pasu največ okoli -114,1 dBm. Izračunani nivoji šuma, ki so odvisni

od usmerjenosti antene, pasovne širine signala, temperature in prisotnosti drugih signalov

v okolici zemeljske postaje, so nižji, in sicer okoli -140 dBm za UHF nivo in okoli -146

dBm za VHF nivo. Tako sprejet šum, kot sprejet signal, bo nato ojačan na signalni poti do

oddajno-sprejemnega modula, tako da bo vhod v ta modul signal z dovolj visoko energijo.

Na signalni poti bo prav tako prišlo do dodatnih šumov, vendar bo razmerje signal/šum na

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koncu še vedno 16,4 dB za UHF komunikacijo in 11,7 dB za S-band komunikacijo, kar je

več od zahtevanega 5-7 dB na sprejemniku.

V smeri proti satelitu so signali na Zemlji pred oddajo s pomočjo močnostnih ojačevalcev

ojačani do te mere, da jih na satelitu po sprejemu ni potrebno ojačevati. Signali, ki

dosežejo sprejemnik na satelitu, bodo imeli moči med -110 dBm in -85 dBm v S-band pasu,

ter ter med -115 dBm in -85 dBm v VHF frekvenčnem pasu. Oboje moči na sredini

območja, ki ga sprejemnik zmore sprejemati.

Ojačanja v zemeljski postaji bodo morala biti šibkosti Trisat signala na površju Zemlje

primerno velika. Poleg ojačanja sprejemno-oddajnih anten, ki so na UHF frekvenčnem

pasu 16,4 dB, bo za ustrezno moč signala na vhodu v sprejemni sistem potrebnih 90 dB

ojačanja. Na S-band pasu so antene bolj usmerjene in imajo 35,4 dB dobitka, prav tako

ima dobitek 30 dB pretvornik na nižje frekvence („downconverter“). Na sprejemu S-band

signalov je tako potrebnih dodatno 55 dB ojačanja. Pri oddaji v S-band pasu oddajnik

oddaja z močjo 17 dBm, pretvornik na višje frekvence („upconverter“) ima 17 dB

ojačanja, dodatni močnostni ojačevalec pa 31, s čimer so dosežene potrebne izhodne moči.

Na VHF pasu zadošča močnostni ojačevalec s 36 dB ojačanjem. Efektivne (usmerjene)

izsevane moči iz antene na Zemlji so 85,92 dBm za S-band in 64,31 za VHF pas, kar

zadošča za dosego zahtevanih moči signalov v okolici satelita pri danem satelitskem

sistemu.

BPSK MODEL

Posvetili smo se tudi izvedbi oddajnika in sprejemnika, cilj pa je bila izdelava modela

BPSK demodulatorja za S-band območje. Oddajnik za BPSK je relativno preprost, saj gre

za stikalo ki preklaplja med dvema viroma za 180° zamaknjenega sinusnega signala

izbrane moči. Sprejemnik na drugi strani mora izvajati Avtomatsko kontrolo dobitka AGC

(„Automatic Gain Control“), obnovo časa, obnovo nosilca in poenotenje kanala

(„Channel equalization“) v primeru, da je sprejetih več zakasnjenih in skaliranih različic

signala, kar je posledica potovanja signala skozi medij.

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Izdelali smo tudi model BPSK demodulatorja. BPSK modulacija je najpreprostejša in

najrobustnejša PSK modulacija, demodulator pa smo razvili na način, ki omogoča

kasnejšo nadgradnjo na PSK višjega nivoja. Demodulator je lahko koherenten ali

nekoherenten oziroma diferenčni demodulator. Koherenten BPSK demodulator zahteva

stalen referenčni signal (val) s katerim primerja prejet informacijski signal, medtem ko

diferenčni BPSK demodulator zaznava zgolj spremembe med nivoji, na osnovi katerih

izlušči zaporedje bitov iz signala. Slabost diferenčnega demodulatorja je večja verjetnost

napak pri demodulacijah.

V našem primeru bo zaradi dosegljivosti opreme na satelitu za komunikacijo izbrana

diferenčna BPSK modulacija. Pregledali smo različne možne implementacije diferenčnih

demodulatorjev ter se odločili za najpogosteje uporabljano implementacijo s Costasovo

zanko. Prednost te implementacije je, da jo je mogoče nadgraditi na PSK višjega reda z

uporabo istih principov, kot so uporabljeni za demodulacijo BPSK signala.

Izdelali smo Matlab model BPSK demodulatorja, prilagojenega za signal s pričakovanimi

karakteristikami (pasovno širino, bitno hitrostjo, frekvenco). Prav tako smo izdelali model

modulatorja in model kanala, ter modele med seboj povezali. Na ta način smo modelirali

celotno pot BPSK signala od oddajnega do sprejemnega sistema.

Ker je izvedba BPSK demodulatorja bila predvidena za FPGA implementacijo v Zemeljski

postaji, smo naredili tudi korak k prototipiranju demodulatorja v FPGA. Uporabili smo

Xilinx FPGA in orodje Xilinx System Generator. Orodje Xilinx System Generator deluje v

Matlab/Simulink okolju, v njem pa je mogoče izvesti FPGA funkcionalnost na visokem

nivoju, s povezavo Xilinx pripravljenih funkcijskih blokov (v praksi gre za IP jedra z

grafično predstavitvijo v Simulink okolju). Nekateri bloki so funkcionalno identični tistim v

standardnem Matlab okolju, nekateri pa imajo drugačne funkcionalnosti. Orodje omogoča,

da programiramo in v zasnovo vstavimo tudi svoje bloke. Pri prenosu abstraktnega Matlab

modela v Xilinx System Generator se hitro srečamo z omejitvami strojne opreme, kot je

frekvenca delovanja sistema, natančnost in bitna širina registrov in povezav, izbrani

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številski formati v registrih in na povezavah in podobno. Prvotno implementacijo v

Matlab/Simulink okolju smo tako morali spremeniti in optimizirati (predvsem filtre), da jo

je bilo mogoče izvesti v Xilinx FPGA.

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BPSK Modulated Satellite Communication System Model

Keywords: digital, phase, modulation, satellite, communication, model, budget, station

UDK: 621.39:004.77(043.2)

Abstract

In thesis, communication system for Trisat satellite mission is proposed and modelled,

based on analysis of the space mission's parameters. End-to-end communication between

Ground station computer and Space station computer is proposed from physical to

application layers, with in-depth analysis of link budget and physical signal properties.

Also BPSK demodulator for Ground station in FPGA is analyzed in-depth, created and

modelled, and Ground station design that allows multiple computers to connect to satellite

over IP transparently is proposed.

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Contents1 Introduction.........................................................................................................................12 Task exploration..................................................................................................................33 CubeSat project presentation...............................................................................................54 FERI CubeSat project..........................................................................................................6

4.1 Expected orbit..............................................................................................................64.2 Frequencies..................................................................................................................7

4.2.1 Amateur Radio Frequencies.................................................................................74.2.2 Selected frequencies for communications...........................................................9

5 System components...........................................................................................................115.1 Satellite communications system...............................................................................12

5.1.1 Satellite antennas...............................................................................................125.2 Path between Satellite and Ground station................................................................135.3 Ground station...........................................................................................................13

5.3.1 Ground station antennas.....................................................................................145.3.2 Receiving system design for S-band..................................................................155.3.3 Transmitting system design for S-band..............................................................175.3.4 Computer clients and local area connection......................................................17

6 Link budget........................................................................................................................206.1 Radio Wave Propagation...........................................................................................23

6.1.1 Atmospheric losses ............................................................................................246.1.2 Free space loss...................................................................................................246.1.3 Doppler effect....................................................................................................256.1.4 Ionospheric losses..............................................................................................25

6.2 Link attenuation.........................................................................................................256.3 System noise temperature..........................................................................................266.4 Antenna noise temperature........................................................................................266.5 Antenna-receiver cable noise temperature.................................................................286.6 receiver noise.............................................................................................................286.7 Carrier to Noise ratio at receiver output....................................................................286.8 Calculations for Trisat...............................................................................................29

6.8.1 Link budget calculation......................................................................................296.8.2 Satellite system and signal levels.......................................................................316.8.3 Calculation results for Satellite system..............................................................31

6.9 Ground station receiver.............................................................................................336.9.1 Receiving system attenuation............................................................................336.9.2 Received noise at the antenna............................................................................346.9.3 System noise......................................................................................................356.9.4 Scheme of Ground stations receivers and transmitters analogue signals path...376.9.5 Calculation results for Ground station receivers................................................38

6.10 Ground station transmitters.....................................................................................386.10.1 Calculation results for Ground station transmitters.........................................39

7 Modulation........................................................................................................................407.1 Why do we need modulation.....................................................................................407.2 Which modulation to choose.....................................................................................41

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7.3 Symbols, bits and bauds............................................................................................428 BPSK modulation..............................................................................................................43

8.1 PSK – Phase-shift keying..........................................................................................438.1.1 Modulator and demodulator relationship...........................................................43

8.2 BPSK – Binary PSK..................................................................................................438.3 Differential coding.....................................................................................................47

9 NRZ encoding...................................................................................................................4810 Overview of tasks at the receiver....................................................................................5011 Demodulator implementations........................................................................................51

11.1 Coherent demodulator implementation...................................................................5111.2 Carrier recovery for coherent demodulation............................................................52

11.2.1 Phase-Locked Loop..........................................................................................53PLL model..............................................................................................................54Second-order PLL..................................................................................................55Third-order PLL.....................................................................................................55Digital Phase-Locked Loops..................................................................................56Digital Tanlock Loop..............................................................................................58

11.2.2 Non-Data-Aided methods................................................................................59Multiply-filter-divide.............................................................................................59Costas loop.............................................................................................................59Squaring loop.........................................................................................................61Open-loop carrier recovery structures....................................................................61Phase ambiguity.....................................................................................................64

11.3 Differential PSK......................................................................................................6512 Bit Error Rate for BPSK modulation..............................................................................6813 Telemetry synchronization and channel coding..............................................................73

13.1 Error-control coding................................................................................................7413.2 Frame validation......................................................................................................7413.3 Synchronization.......................................................................................................7513.4 Pseudo-randomizing................................................................................................7513.5 Organization of Synchronization and Channel Coding Sublayer of the receiver....75

14 System specifications......................................................................................................7714.1 Signals specifications..............................................................................................7714.2 Downconversion specifications...............................................................................77

15 Costas loop parameters in practice..................................................................................7916 BPSK system model........................................................................................................81

16.1 BPSK modulator block............................................................................................8216.2 Channel block..........................................................................................................8316.3 Downconversion block............................................................................................8616.4 BPSK demodulator block........................................................................................8716.5 Model in practice.....................................................................................................8716.6 Step towards Implementation in hardware..............................................................89

17 Conclusion.......................................................................................................................91Bibliography.........................................................................................................................93Appendixes...........................................................................................................................98

A Link budget calculations..............................................................................................98

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A.1 Ground station part – downlink...........................................................................98A.2 System noise calculations....................................................................................99A.3 Ground station part – uplink..............................................................................101A.4 Satellite part.......................................................................................................102A.5 Path loss calculations.........................................................................................104

B Xilinx System Generator prototype of demodulator..................................................106B.1 System Generator model specifications.............................................................106B.2 System Generator model....................................................................................106B.3 System Generator model of demodulator...........................................................107B.4 System Generator model of NCO......................................................................108

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Figures indexFigure 6.1: Orbital parameters. [27].......................................................................................7Figure 7.1: High-level system overview...............................................................................11Figure 7.2: Ground station transceiver hardware design......................................................16Figure 7.3: Server application design proposal, improved design, based on [21]...............18Figure 8.1: Wireless communication system [22]................................................................21Figure 8.2: Antenna noise power. [22].................................................................................27Figure 8.3: Analogue signals path in Ground station...........................................................37Figure 10.1: BPSK constellation..........................................................................................44Figure 10.2: Carrier signal, baseband modulation signal and resulting BPSK modulation. [32].......................................................................................................................................45Figure 10.3: BPSK frequency spectrum (symbol frequency: 600 Hz) [32].........................46Figure 11.1: NRZ encoding example [34]............................................................................48Figure 13.1: Coherent demodulator......................................................................................52Figure 13.2: Simple PLL model. [37]..................................................................................54Figure 13.3: Simplified DPLL block diagram. [38].............................................................56Figure 13.4: FF-DPLL block diagram [38]..........................................................................57Figure 13.5: LL-DPLL block diagram [38]..........................................................................57Figure 13.6: ZC-DPLL block diagram [38].........................................................................58Figure 13.7: Digital tanlock loop [48]..................................................................................58Figure 13.8: Costas loop [40]...............................................................................................60Figure 13.9: Squaring loop structure [37]............................................................................61Figure 13.10: First-order ML phase estimator [37]..............................................................63Figure 13.11: Modified First-order ML phase estimator [37]..............................................63Figure 13.12: Second-order ML phase estimator [37].........................................................64Figure 13.13: Differential PSK modulator and demodulator [40].......................................65Figure 13.14: Differential BPSK demodulator [42].............................................................65Figure 13.15: BPSK demodulator wit EPLD Delay Line [42].............................................67Figure 13.16: Noise in the channel.......................................................................................69Figure 13.17: Treshold, probability of received signals and error probability [44].............70Figure 15.1: CCSDS protocols, ISO/OSI model and our FPGA-based demodulator functions. Based on Figure2-1 in [45].................................................................................74Figure 15.2: Synchronization and Channel Coding sublayer functions [45].......................76Figure 17.1: Digital Costas loop model, based on [46]........................................................80Figure 18.1: BSPK system model in Simulink.....................................................................82Figure 18.2: BPSK modulator Simulink model...................................................................83Figure 18.3: Channel simulation block................................................................................83Figure 18.4: Imaginary number representation....................................................................85Figure 18.5: Downconversion block in Simulink................................................................87Figure 18.6: Original recovered bit sequence in system without signal impairment in the channel..................................................................................................................................88Figure 18.7: Original and recovered bit sequence with signal impairments in the channel and AWGN noise with SNR of 11 dB..................................................................................89

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ABBREVIATIONS LISTFPGA Field-Programmable Gate ArrayBPSK Binary Phase Shift KeyingVHF Very High FrequencyUHF Ultra High FrequencySPI Serial Peripheral InterfaceSNMP Simple Network Management ProtocolUDP User Datagram ProtocolTCP Transmission Control ProtocolNI USRP National Instruments Universal Software Radio PeripheralRF Radio FrequencyAGC Automatic Gain ControlISO/OSI International Standards Organization / Open Systems InterconnectionPSK Phase-Shift KeyingAFSK Audio Frequency Shift KeyingQPSK Quadrature Phase-Shift KeyingASCII American Standard Code for Information InterchangeIP Internet ProtocolRHCP Right Hand Circular PolarizationESMO European Student Moon OrbiterDAC Digital to Analog Converter/ConversionADC Analog to Digital Converter/ConversionANTLR ANother Tool For Language RecognitionEIRP Equivalent Isotropically Radiated PowerSNR Signal to Noise RatioSDR Software Defined RadioLNA Low-Noise AmplifierNRZ Non-Return-to-ZeroAD Analog to Digital

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DA Digital to AnalogPLL Phase Locked LoopVCO Voltage-Controlled OscillatorVLSI Very-Large-Scale IntegrationDPLL Digital Phase Locked LoopLPF Low Pass FilterAWGN Additive White Gaussian NoiseELPD Electrically Programmable Logical DeviceBER Bit Error RateLDPC Low-Density Parity-CheckITAR International Traffic in Arms RegulationsARRL American Radio Relay LeagueITU International Telecommunications UnionIARU International Amateur Radio Union

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BPSK MODULATED SATELLITE COMMUNICATION SYSTEM MODEL Page 1

1 INTRODUCTION

Since October 4, 1957, when Soviet Union launched the first artificial satellite into orbit

around Earth [1], artificial satellites are being launched and used for different purposes,

from scientific purposes, communication, broadcasting, navigation aid and more.

One of the satellites which will be orbiting around Earth will be also Trisat, a satellite

developed by University of Maribor and its industrial partners.

The goal of this Master thesis is to analyze end to end satellite communication system

including communication media, and propose design of ground communications station for

both directions of communication [2].

Generally, the parameters in Earth-space communications and their influence on

communications link performance are known, but must be calculated separately for every

mission according to misson's objectives, orbital parameters, available frequencies for

communication, and available technology on the spacecraft and on ground station. In this

Master thesis, complete communications link is analyzed and ground station's design is

proposed for Trisat mission.

Also, a design of a ground station's receiver, built around FPGA and ARM processor, is

proposed, including Matlab's model of demodulator which is to be implemented in FPGA.

The proposed design allows multiple host computers to connect to a receiver by using IP

communication.

In the thesis, we first clarify the details about satellite, its orbital parameters and selected

frequencies for communication between ground station and spacecraft.

Then we move on to description of the whole communications system from transmitter and

receiver on Earth, over the signals path, to receiver and transmitter on satellite. In this part

of the thesis, we outline the design of all of the components of the system. The description


consists of details about the involved components and the path, traveled by the signals.

Also, the standards and protocols, used throughout the system, are described there. As

lowest layers of communications (physical and modulation layer) are more specific for

every mission because of specific physical properties of the mission (like orbit,

frequencies, data rate requirements), more detailed research on physical level and

modulation level follows.

Therefore in the next chapters, link budget is considered and calculated, followed by

overview of modulations, particularly BPSK modulation. Also a BPSK demodulator model

for S-band frequencies is described.

There is also some theory about channel encoding for space missions, as it is the theme

that also composes final performance of the link in terms of effective transmission data

rate.

The conclusion, based on results, can be found in the conclusion chapter.


2 TASK EXPLORATION

Digital communications system can be described as a composition of abstract layers by

using an ISO Open Systems Interconnection (OSI) model. ISO/OSI model consists of 7

layers: Physical layer, Data link layer, Network layer, Transport layer, Session layer,

Presentation layer and Application layer. Each of the layers does its own function and

serves the layer above. Application software on both ends of communication finally

communicates through application layer [3].

In our satellite system, communication occurs only between two endpoints, the Ground

station and the Space station. Both stations should have full stacks (from Physical layer to

Application layer) implemented for successful communication.

Ground station will be divided to transmitting and receiving module, the necessary

components for signal attenuation and manipulation, and server computers. Transmitting

and receiving module will will consist of FPGA module, ARM processor, Ethernet

communication for link with computer over network, and necessary components for RF

communication towards satellite. Design of Ground station system is to be proposed.

To propose design of Ground station's RF part, we need to know the properties of signals

that will be received and transmitted. Those are based on physical characteristics of

satellite communication system, like orbit and frequency. Based on those characteristics,

we can estimate losses on the path between Space station and Ground station, and calculate

the Link budget for the whole system, which is another task of this thesis.

Further adaptation of low-layer communication to the task will be then simulated by


developing a modulator/demodulator model for one of the used frequency bands, in our

case for S-band. Modulation is going to be phase modulation (Phase-Shift Keying), namely

BPSK (Binary Phase-Shift Keying).

Because of the fact that if higher PSK will be used, it is possible to upgrade BPSK

implementation in FPGA to higher order PSK „relatively simply“ if we use the right

design, the goal is to implement a BPSK demodulator first.

The FPGA will also perform basic symbol flow and frame decoding.

The second phase will be decoding of frames which define packets.

As the result, the whole communication system will be proposed, with more in-depth

research of Physical layer.


3 CUBESAT PROJECT PRESENTATION

The CubeSat Project is an international collaboration project with more than 40

universities, high schools and private firms involved. Resources are available by

communicating with other developers, as well as there are organized CubeSat workshops.

A CubeSat is a satellite with a 10 cm cube shape and a mass up to 1.33 kg. Also, tripple

CubeSat can be developed, with 10x10x30 cm shape and mass up to 4 kg. Usually,

CubeSats are placed into a rocket as a secondary payload.

The project began in 1999, when California Polytechnic State University (Cal Poly) and

Stanford University developed specifications for satellites to allow space exploration for

universities worldwide through a unified program.

A CubeSat standard is designed to provide developers with necessary guidelines to

interface wih Poly Picosatellite Orbital Deployer (P-POD), the module for satellite

deployment into orbit developed at Cal Poly. The standard describes outer dimensions of

the satellite, recommended materials, restrictions and describes schedules that need to be

followed during integration and launch.

Cal Poly is involved in obtaining required documentation, conforming to ITAR

(International Traffic in Arms Regulations) and organizing final delivery of integrated P-

PODs to launch site [4].

All that is necessary to participate in CubeSat program is to design and build a satellite

according to that standard [5].

Since 2012, there have been already 11 successful launches [6] and there are plans for

further launches around the globe.


4 FERI CUBESAT PROJECT

The primary goal of the satellite is to cover educational, scientific and technological

aspects of space missions.

The satellite will be a 3U CubeSat spacecraft with mass of 3.75 kg and will operate in Low

Earth Orbit.

Trisat will take visible and near-infrared images of Earth surfaces, perform spectral

analysis of sun's visible to infrared radiation, and fly autonomously with on-board data

handling [2].

4.1 EXPECTED ORBIT

The orbit of the satellite is not yet totally defined. The expected Height of Apogee of the

orbit is 1000 km, the expected Height of Perigee is 600 km. Expected elevation angle is 45

degrees. The parameters of the orbit are shown in Figure 6.1.


4.2 FREQUENCIES

Amateur Radio frequencies will be used. The communication between Ground station and

Satellite will occur on VHF, UHF and S-band frequency bands. S-band will be used for

higher data rate (capability of satellite transciever is up do 2 Mbps) transmission of

payload data, while VHF (uplink) and UHF (downlink) will be used for lower data rate

transmission (1200 to 9600 baud) of telecommands. Data rates were selected based on

transceiving equipment which will be installed on the satellite.

4.2.1 AMATEUR RADIO FREQUENCIES

Cubesat satellites are going to use Amateur Radio frequency bands for communication

Figure 6.1: Orbital parameters. [27]


between Earth and satellite. The Cubesat Developers Documents [6] suggests us to follow

the Part 97 of Federal Regulatory Title 47 – Telecommunications ([7]), which is available

on ARRL1 website.

The frequency band regulations in [8] are generally aligned with international regulations

for Amateur Radio usage, ordered by International Telecommunication Union (ITU).

The frequencies for satellite and space craft communications are allocated separately from

general use radio amateur bands. The ITU's rules state that all amateur radio operations

may occur withing 50 kilometers of Earth's surface, while Amateur Satellite Service rules

regulate communication with space crafts and satellites. This is partly because of the fact

that there are differences in allowed frequency bands for Amateur Radio usage in different

regions of Earth, while those space crafts circle around the planet and so should have the

frequency allocations, compatible with all of the regional regulations.

In subpart C - Special Operations of [9], allowed frequency bands for amateur radio

communication with space stations are defined. As Cubesat satellite is an example of a

space station, there are following bands and segments in which amateur radio stations are

authorized to transmit:

"(1) The 17 m, 15 m, 12 m and 10 m bands, 6 mm, 4 mm, 2 mm and 1 mm bands; and

(2) The 7.0-7.1 MHz, 14.00-14.25 MHz, 144-146 MHz, 435-438 MHz, 1260-1270 MHz

and 2400-2450 MHz, 3.40-3.41 GHz, 5.83-5.85 GHz, 10.45-10.50 GHz and 24.00-24.05

GHz segments." [10]

Ground stations are authorized to transmit in mostly similar frequency segments with some

differences in bands between 5-6 GHz:

"(1) The 17 m, 15 m, 12 m and 10 m bands, 6 mm, 4 mm, 2 mm and 1 mm bands; and

(2) The 7.0-7.1 MHz, 14.00-14.25 MHz, 144-146 MHz, 435-438 MHz, 1260-1270 MHz

and 2400-2450 MHz, 3.40-3.41 GHz, 5.65-5.67 GHz, 10.45-10.50 GHz and 24.00-24.05

1 The national association for amateur radio, United States of America


GHz segments." [10]

Those frequencies are allowed for Space Telecommand stations operation.

IARU defines the following frequency allocations to amateur-satellite service, which is

shared with other services:

"435 - 438 MHz, 1 260 - 1 270 MHz, 2 400 - 2 450 MHz, 3 400 - 3 410 MHz (available in

Regions 2 and 3 only), 5 650 - 5 670 MHz, 5 830-5 850 MHz, 10.45 - 10.5 GHz, 76 - 81

GHz, 144 - 149 GHz, and 241 - 248 GHz. " [11]

Because the satellite communication is happening in Amateur Radio bands, all of the

communication must be open, therefore not encrypted in any way. Because of the fact that

digital communications use different codes and formats, all of this must be made publicly

available to meet the plain language requirement of Amateur Radio communications.

The operations on the communications link to the satellite must not serve commercial

proposes or proposes of getting finances [11].

The frequency allocation for particular satellite is in the end done by International Radio

Amateur Union (IARU) and should be requested by sending a form, available at [12], to

IARU. In the process, due to the fact that some frequencies may be already used, the

specifications may change so the form should be sent as much in advance as possible.

In the crowded parts of the orbit, the satellites can be relatively close to each other and

there may also be frequency discrimination between neighbouring satellites. For example,

in the crowded parts of geostationary orbit, the satellite spacing is about 2°.

4.2.2 SELECTED FREQUENCIES FOR COMMUNICATIONS

Used frequency ranges are UHF, VHF and S-band. UHF and VHF communications are not

meant to be well pointed as they have to function also when the satellite will not be pointed

in desired direction towards the Ground station. S-band communications will be used when


satellite's S-band antennas will be pointed towards Ground station.

The frequencies are not yet totally defined, but desired frequencies are the following:

in VHF band: in the band from 144 to 146 MHz, preferred 145.800 MHz to 146.000 MHz.

In UHF band: 435 MHz do 438 MHz.

In S-band: 2.400 GHz to 2.450 GHz, preferred 2.400 GHz to 2.403 GHz.

Those frequency bands are Amateur Radio frequency bands.

Exact frequencies are not known yet, so we used expected frequencies in radioamateur

ranges (described in 4.2.1 Amateur Radio Frequencies) in VHF range, frequency used for

calculations was 145,8 MHz, in VHF range we used 437,45 MHz and in S-band, we used

frequency of 2228 MHz for calculations.

VHF and UHF communication will occur simultaneously and will present main

communication channel for Telemetry and Telecommands communication. UHF and VHF

communication system will utilize a full duplex, frequency divided communication

channel. 15 kHz bandwidths will be used in VHF and UHF, and selected modulation

scheme is BPSK in UHF and FSK or GFSK in VHF. The communication on UHF and

VHF should be possible disregarding the orientation of the satellite.

S-band communication is a secondary communication channel for higher-speed

communication, when the satellite is aligned – oriented in proper direction with S-band

antenna system on satellite facing S-band antenna system on ground station. 50 MHz

bandwidth is used, and selected modulation scheme is BPSK.


5 SYSTEM COMPONENTS

The communication system consists of multiple components, as shown in Figure 7.1.

In the following sections, the components of the system and their interconnections are

described.

Figure 7.1: High-level system overview.


5.1 SATELLITE COMMUNICATIONS SYSTEM

Satellite's UHF downlink/VHF uplink transceiver is capable of GMSK at 9600 bauds

(symbols per second) and AFSK at 1200 bauds. It is able to transmit on frequencies

between 420 MHz and 450 MHz with output RF power between 27 and 33 dBm.

Its reception capabilities consist of Noise figure < 1.5 dB, dynamic range between -120

dBm and -70 dBm, and frequency range between 130 MHz and 150 MHz. [14]

The S-band transmitter supports amateur radio frequency bands on S-band, data rates of up

to 2 Mbps with QPSK and output powers between 21 dB and 30 dBm. [15]

5.1.1 SATELLITE ANTENNAS

On satellite, there will be 2 kinds of antenna systems: one for UHF and VHF, and one for

S-band.

UHF and VHF antennas will be dipole antennas. Because the basic communication will

occur on UHF and VHF frequencies, the antennas for this communication need wide

opening angle. That way, satellite can communicate with ground station disregarding

satellite's orientation. Monopole antennas will be used and will have low attenuation,

almost 0 dBi.

S-band antenna will be patch antenna with about 50 MHz of RF bandwidth and

approximately 6 dBi gain. Frequency used can be between 2200 MHz and 2300 MHz, and

its opening angle is approximately 85 degrees. This antenna will be used for pointed

communications, therefore for communication between satellite and ground station after

the satellite will be pointed in the right direction so the antenna will point towards the

ground station on Earth. The final antenna design may differ slightly because of fine tuning

to final confirmed and selected frequencies. [16]


5.2 PATH BETWEEN SATELLITE AND GROUND STATION

There will be BPSK radio communication on selected frequencies (described in 4.2.2

Selected frequencies for communications). Link budget and factors that influence the

signal on the path between Satellite and Ground station have been researched and are

described to larger extent in 6 Link budget. S-band signal bandwidth will be up to 50 MHz

and UHF and VHF signal bandwidth will be up to 15 kHz.

5.3 GROUND STATION

Structure of the Ground station will be further divided into transmitting/receiving system

and client computers with application software to interface the communication system.

Transceiver handles communication with Space station over space link on one side, and

with server computers over Ethernet network. On each server computer, there is special

application software which allows setting of different parameters of transceiver and

interfacing of satellite.

Communication between Space station and transciever will will consist of low data rate

communications on VHF (for uplink) and UHF (for downlink) bands, and higher data rate

communications on S-band. Low data rate communications will be used for

Telecommands, therefore commands for controlling satellite's basic function. Higher data

rate communications will be used for transmitting larger data from and to satellite (like

images from satellite and software updates to satellite).

Telecommands will be sent from server computer to transceiver system, and from there to

satellite, so the transciever system will need full communication stack for handling this

data. In the same way, low data rate communication will occur in the opposite direction,

therefore from satellite to server computers. Higher reliability and transparency of

communication is achieved that way.


Protocol for sending telecommands will be developed, while on lower level, CCSDS will

be used as described in [17].

Higher data rate communication will flow through transceiver system directly from

sattelite to server computer, without intermediate saving of the data in the transceiver

system. The transceiver system will just re-encapsulate data on lower layers of OSI model,

and will not handle higher layers of data. Lower complexity and higher speed is achieved

that way.

Proposed communication protocol between client computer and transciever is SNMP for

setting values on transceiver's RF components and special ASCII based protocol on top of

TCP/IP for chunks of Telecommands, and UDP encapsulated data stream for high-speed

transmission of data from satellite to server computer.

5.3.1 GROUND STATION ANTENNAS

For S-band communications, a 3 meter dish antenna will be used. Its gain is 35,4 dBi,

while the -3dB angle is only 3,2 degrees. This antenna has RHCP polarization and will be

used for uplink and downlink S-band communications because of its high gain and

practical reasons like proper dimensions to be mounted on our Ground station and

affordability. [18]

For telemetry upload, yagi antenna with 2x9 elements will be used. Antenna is suited for

144 to 146 MHz bands and has 13,1 dBi isotropic gain with 2x20 degrees (H-plane) and

2x23 degrees (E-plane) of aperture angle and is used because it suits given frequency

ranges and physical requirements for antenna. [19]

For telemetry download, 19 element yagi antenna will be used. It is suitable for frequencies

between 430 MHz and 440 MHz, and has an isotropic gain of 16.4 dBi with aperture angle

of 14,8 degrees (H-plane) and 15,7 degrees (E-plane). Radiation patterns can be found in


referenced antenna documentation. [20]

5.3.2 RECEIVING SYSTEM DESIGN FOR S-BAND

In the first phase, NI-USRP 2920 universal software radio peripheral will be used to speed

up the proces of Ground station development. That's why downconversion blocks are used

on the signals reception path. Downconversion will be done from 2000 MHz – 2700 MHz

band down to 167 MHz – 867 MHz band (according to [13], page 33). Most of the

requiremens in Link budget section (6 Link budget) are based on NI-USRP 2920

specifications.

Part of the system was developed preliminary by author of this thesis at University of

Maribor, primarily for ESMO mission and is described in [21]. It included only

adjustments of RF-related component's settings from PC over Ethernet-based IP network,

by using ARM processor and FPGA on-board. The idea of implementation is briefly

presented in this section, while it's upgraded towards fully capable transceiver on

conceptual level, and improved in terms of used protocols to suit Trisat mission .

Hardware design of the recieving system consists of software controlled demodulator,

developed in FPGA, which is accessible over Ethernet IP network. IP stack with basic

software is to be run inside ARM processor on board of receiving system.

There are multiple RF-related components connected to FPGA, which are to be controlled

over the network [21]:

• clock generator

• attenuators

• synthesizer

• double ADC (analogue-to-digital conversion)


• quadrature upconverter

• DAC (digital-to-analogue conversion)

The scheme of the system is shown in Figure 7.2.

Because all of the RF-related components are connected to FPGA, the FPGA needs to

route the communication between ARM processor and components. The ARM processor

can then control RF components based on the data from IP network.

Each of the components therefore needs its own adress on the bus and FPGA must be able

to route the data between bus to ARM and bus to specific component, based on the address.

Besides, FPGA must be able to initialize the components and should hold the design and

functionality of demodulator, described in 16 BPSK system model.

There are two Ethernet interfaces proposed on the board, one connected to ARM processor

for communication with PC where full IP stack is needed, and the other, connected directly

Figure 7.2: Ground station transceiver hardware design.


to FPGA, for high-speed payload data from satellite, where only UDP encapsulation of

data stream, which is exiting demodulator, needs to be done.

ARM handles IP communication, with SNMP protocol parsing, SNMP responses, and data

setting on RF components over FPGA switch.

Design from the RF power point of view is descibed in 6.8.1 Link budget calculation.

5.3.3 TRANSMITTING SYSTEM DESIGN FOR S-BAND

BPSK transmitting system consists of two sine wave generators with the same frequency

and 180° phase difference. Switching between generators is done, based on the bit stream

that needs to be modulated. The signal is then attenuated inside the transmitting system in a

way that it meets RF power specifications, described in 6.8.1 Link budget calculation.

In the first phase, NI-USRP 2920 universal software radio peripheral will be used to speed

up the proces of Ground station development. That's why upconversion blocks are used on

the signals transmission path. Upconversion will be done from 597 MHz – 797 MHz band

up to 2300 MHz – 2500 MHz band (according to [13], page 33). Most of the requiremens

in Link budget section (6 Link budget) are based on NI-USRP 2920 specifications.

5.3.4 COMPUTER CLIENTS AND LOCAL AREA CONNECTION

Local area connection is done over IP-based Ethernet network. Protocol to be used by

application to set and read values in RF components is SNMP, because of its simplicity and

because it suits well to the purpose of simple transport of values over network.

The data stream from satellite on S-band should be just encapsulated into UDP and sent to

the server computer, which should then deal with the data stream.

Main goal of the application is to control BPSK transceiver parameters. Besides, it may


also handle telecommands for satellite. Data, encapsulated in UDP and transmitted from

satellite, should be handled with another application on server computer (which may be

also another server computer). Application should also allow saving and loading files with

parameters and telecommands for further use.

The proposed design of the application on the server computer is shown on Figure 7.3.

Application is proposed to be developed in Java because of readily available interfaces and

libraries for network communication and tools for other purposes (like input parsing), and

easy portability to different platforms for desktop computers. It consists of the following

components [21]:

User interface deals with user interaction. Its main component is window for text-based

input of key-value pairs, with which user can assign values for different parameters to be

set on transceiver, or telecommands to be set to satellite.

Alternatively, telecommands can be the input in user interface.

Formatter of user input can be built with ANTLR and should check if input data is

correct, parse the input data, and pass it to Protocol automate.

Figure 7.3: Server application design proposal, improved design, based on [21].


Protocol automate handles SNMP protocol communication on data and protocol level

(abstract). Encapsulator/decapuslator is the module that takes care of packeting data into

SNMP protocol units. Those units are then sent over network. During the reception of the

protocol unit from transceiver, the path through application is reverse.

Instead of developing Protocol automate, Encapsulator/decapsulator and Network

handling modules, some SNMP library can be used.

For telecommands sending, alternative ASCII based protocol can be used on top of TCP/IP

stack for sending chunks of telecommands over IP network to transceiver.

Modules for program settings and file handling are acting as support modules for basic

functionality of application, as described before.


6 LINK BUDGET

There are multiple factors that affect the signal which enters the demodulator. Besides the

factors that affect the signal on its path through space, the receiver and transmitter

parameters are very important, as well as the signals from other satellites and sources add

to the received signal.

In receiver and transmitter design, the parameters of antennas play important role in

conjunction with used power for signal transmission.

The composition of all gains and losses on the path from transmitter, through the medium

that the signal traverses, to the receiver in a telecommunication system, is called a link

budget and can be calculated as:

received power (dBm) = Transmitter power (dBm) + Gains (dB) – Losses (dB).

In the following section, we will describe some factors that influence the link budget.

The system is shown on Figure 8.1


On Figure 8.1:

• Tx represents the transmitting side of the link.

• GT represents gain of transmitting antenna and PT represents transmitted power.

• Rx represents the receiveing side of the link.

• GR represents gain of transmitting antenna and PR represents transmitted power.

• R is the distance between the two antennas.

PTGT is called the Effective Isotropic Radiated Power (EIRP) and gives a measure of the

power flux. It is usually given in units of dBW, which means [22]:

EIRP[dBW] = 10log10(PTGT)

For a satellite, contours of constant EIRP can be plotted on Earth's surface.

If Tx transmits as a sphere, then power in the sphere at the distance of the recieving

antenna is (8.1):

Figure 8.1: Wireless communication system [22].


P=PT⋅G4Π⋅R2 (8.1)

Because of the finite effective area of the antenna, the power at the receiver antenna is

(8.2):

(8.2)

where Aeff represents the receive antenna effective area.

Because antenna gain can be represented as (8.3):

(8.3)

where λ is signal wavelength in meters. The power at the receiver can be represented as

(8.4):

(8.4)

However, because of antenna efficiency (some power is lost in the antenna feed structure

and connections to the receiver), polarization mismatches of Tx and Rx antennas,

misalignments of Rx and Tx antennas, atmospheric absorption and other effects, described

in 6.1 Radio Wave Propagation, the corrections must be added to calculation of PR. These

corrections can be included through additional loss factor L, so the equation becomes (8.5):


(8.5)

Often, the link budget calculations are carried out using powers measured in dBW,

therefore the powers measured relative to a 1 watt reference power [22] (8.6):

(8.6)

then the equation for PR with additional loss factor L, calculations looks like (8.7):

(8.7)

where typically, L is about 5dB. [22]

6.1 RADIO WAVE PROPAGATION

Radio waves, propagating through Space from satellite to ground station on Earth, are

affected by different factors on the way. Those include atmospheric issues, rain attenuation

and ionosphere losses, among other possible factors. Also the Sun and the Earth are two

RF sources.

How those factors affect the radio waves strongly depends on the frequency of the signal

and there is in general the greatest difference between signals above 10 GHz and signals

with frequencies between 1 GHz and 10 GHz.

The frequency range below 10 GHz is already extensively used by terrestrial, as well as

existing satellite services. This means that even though lower frequencies are in general

less sensitive to attenuation, the potential for interference with terrestrial links can present


problems on certain locations. Above 10 GHz, the rain attenuation increases, but there is

less chances for interference with other services.

Within the range of 1 to 10 GHz, there are bands with higher absorption because of

atmospheric components, like water vapor and oxygen. [22]

6.1.1 ATMOSPHERIC LOSSES

• Beam-spreading loss is the result of the signal spreading through Earth's

atmosphere.

• Scintillation loss is the result from rapid variations in signal's amplitude and phase

because of the changes in the refractive index of the Earth's atmosphere

• Polarization loss is the result from rotation of polarization of the signal, as it

passes through the Earth's atmosphere

• Rayleigh Fading is the result from interference, which occurs because the same

signal arrives over many different paths, which results in out-of-phase components

at the receiver.

• Weather losses is the attenuation of the signals which happens because of

hydrometers in the atmosphere. Rain attenuation is predominant loss element at

frequencies below 60 GHz. [23]

6.1.2 FREE SPACE LOSS

Free Space Loss is the major loss of the signals that are traveling between Earth and

satellite. It is inversely proportional to the square of the distance the signal travels and to

the square of the frequency used. It varies with the frequency. [23]


6.1.3 DOPPLER EFFECT

Doppler effect is the change of frequency of the electromagnetic signal, which is caused by

the relative speed of the satellite in relation to Earth station. When the satellite or Earth

station is moving quickly, it is important to consider the Doppler effect.

When orbital parameters of the satellite are known, Doppler shift can be used to determine

the position of Earth station, as well as when Earth station position is known, Doppler shift

can be used to estimate the orbital parameters of the satellite. [23]

6.1.4 IONOSPHERIC LOSSES

Ionospheric effect may be encountered at frequencies around 1.5 and 2.5 GHz, due to

scintillation. The magnitude of ionospheric losses varies considerably with the time of the

day and the sunspot activity level. [23]

6.2 LINK ATTENUATION

The link attenuation α in dB is given as the ratio between received and transmitted power,

which therefore translates to the function of distance and antenna gains [22] (8.8):

(8.8)


6.3 SYSTEM NOISE TEMPERATURE

For satisfying communication, there must be large enough signal at the receiver, as well as

high enough Signal to Noise ratio (SNR). SNR for TV reception, prescribed with

international regulations, SNR >= 47dB. However, our system will be much smaller and

will have lower power capabilities, so we will have to count on lower SNR.

The basic quality of a link is expressed in terms of carrier to noise ratio (C/N ratio), where

C is the power of unmodulated carrier and N is the power of noise. The signal to noise ratio

of a modulated information signal depends on C/N ratio and the type of modulation used.

The noise power N is specified by the system noise temperature Ts, which is made up from

the following:

• antenna noise TA

• antenna-receiver connection (a cable)

• receiver noise TR

The noise power in watts is in each case calculated from noise temperature, which must be

in degrees K, using the general relationship (8.9):

Pnoise=k⋅T⋅B (8.9)

where k is the Boltzmann's constant:

and B is the bandwidth. [22]

6.4 ANTENNA NOISE TEMPERATURE

The reason for antenna noise is the energy, which is fed to the antenna by radiation sources


other than receiving satellite, like other satellites, stars and galaxies. Also because of the

atmosphere's resistance, some noise power is supplied to antenna.

The output noise power will therefore depend on positions and temperatures of other

radiation sources, and on the gain and polar radiation pattern of the antenna.

For ground station antenna, therefore antenna pointing to the sky, the noise power is

represented by the sky temperature Tsky (originating in the atmosphere) and the Earth

temperature Tearth. Different elevation angles of the antenna are the reason for different sky

temperature; the lower the elevation angle, the longer is the path that the signal traverses

through atmosphere. Therefore low elevation angles, for example below 10°, are usually

avoided. Figure 8.2 shows the function of sky noise over different frequency ranges and

with different elevation angles. [22]

Figure 8.2: Antenna noise power. [22]


6.5 ANTENNA-RECEIVER CABLE NOISE TEMPERATURE

The cable that connects antenna to the receiver has a temperature TC. Let's denote insertion

loss as L, then effective noise temperature in cable is (8.10):

T C=T 0⋅( L−1) (8.10)

where TB is temperature. [22]

6.6 RECEIVER NOISE

In the receiver, noise includes contributions from thermal noise, shot noise and possibly

flicker noise. Those noises may arise in the input RF section of the receiver, the mixers and

te IF stages. The total receiver noise can be calculated from the individual contributions

using the formula (8.11):

T R=T 0⋅(F R−1) (8.11)

where FR represents the receiver noise figure. [22]

6.7 CARRIER TO NOISE RATIO AT RECEIVER OUTPUT

We know that received power of the signal at the receiver can be calculated as the

following (8.12):

(8.12)

We also know that noise power at the receiver is (8.13):


N=Pnoise=k⋅T s⋅B

Therefore we can calculate carrier to noise ratio as (8.13):

(8.13)

where PTGT is EIRP of Tx end, B is bandwidth, R is the distance, λ is signal wavelength in

meters, L presents additional loss factor and k is Boltzmann's constant.

(GR/TS) is the receiver figure of merit. For Intelsat ground station for example, it is [22]

(8.14):

(8.14)

6.8 CALCULATIONS FOR TRISAT

6.8.1 LINK BUDGET CALCULATION

Link budget is sum of all gains and losses on the path through atmosphere and space,

between the transmitter and receiver.

In practice, we basically try to conserve as much power as possible, and based on link

budget calculations, calculate the lowest power necessary at the transmitter, to get enough

power at the receiver.

Some of the power is lost in the transmitting station itself, on the path from transmitting

circuit to the antenna. We can say that final transmitted power PT depends on power on the

output of transmitting circuit P0, and the losses and gains in the system.

Because of the fact that losses and gains in the system multiply and divide the level of the


signal, it is practical to pass to the decibel-watt units instead of watts, where (8.15):

PT [dBW ]=10 log10(PT [W ]) (8.15)

What we need to do first is calculation of Effective Isotropic Radiated Power, which is the

effective amount of power, leaving antenna, and accounts also gains and losses in the

antenna, LT and GT:

EIRP=PT+LT+GT (8.16)

When the signal leaves antenna, it propagates first through atmosphere, and then through

space. Some loss occurs in atmosphere, and it depends on frequency and elevation angle of

the antenna pointing to the satellite, and consequently distance which is traveled by the

signal in atmosphere. Atmospheric loss in our case is relatively small, a bit more than 1 dB.

The major loss occurs because of the signal traveling through space. This loss depends on

the distance and frequency, and is called space loss. It is calculated as following (8.16):

(8.16)

where RIP is Received Isotropic Power, λ denotes wavelength, S power per unit area and d

denotes distance of traveled path.

In decibels, path loss is calculated as (8.17):

(8.17)


When the signal reaches the receiving antenna, the antenna's gain is added. Also, the losses

and gain in the receiving station are accounted on the signal's traversal through the

receiving system. [23]

6.8.2 SATELLITE SYSTEM AND SIGNAL LEVELS

The communication system on satellite is generally much more compact than Ground

station system. The received signals from the antenna are fed directly into the

communication circuit on the satellite.

In the Earth-Space communication, we used EIRP values of our Ground station, and

accounted for path loss and satellite antenna gain to get the level of signals which are

entering satellite communication circuit.

We did similar for Space-Earth communication. We used communication circuit output

level range as basis, and accounted for antenna gains and path loss to get signal levels at

the ground station.

In this section of calculation, the path between Ground station's antenna and satellite's

receiving circuit is considered.

6.8.3 CALCULATION RESULTS FOR SATELLITE SYSTEM

We used the following data in our calculation:

EIRP of Ground station

VHF: 23 dBm to 51 dBm

S-band: 49 dBm to 73 dBm

Satellite's communication circuit transmit powers:


S-band: 21 to 30 dBm

UHF: 27 to 33 dBm

Satellite antenna gain at VHF/UHF: 0dBi

Satellite antenna gain at S-band: 6 dBi

Orbit – perigee of 600 km and apogee of 1000 km,

elevation range of 45 degrees,

Communication frequencies

in S-band: 2228 MHz,

in UHF band: 450 MHz,

in VHF band 136 MHz

With gains of selected antennas and performance of transciever systems on the satellite,

sufficient signal levels on Earth surface should be achieved.

Ground Station to Space Station communications will generally exceed the requirements

and the amount of signal energy will be in the middle of the expected input power: -110,5

dBm to -86,5 dBm on S-band and -114,6 dBm to -86,6 dBm on VHF link, where expected

signal range on the receiver is between -120 dBm and -70 dBm.

In the direction from Space station to Ground station, the received signal level will be close

to lower limit of the receiver. Attenuation on Ground station can be achieved in an easier

way, but the signal level in the proximity of Ground station's antenna must still exceed the

level of noise entering Ground station's antenna. This will generally be achieved with

-129,5 dBm signal level on S-band (using highest possible transmit power at the satellite's

transmitter) and -120,2 dBm to -114,2 dBm signal level on UHF band.

Those signal levels are adequate, which can be seen from calculations for Ground station

(can be found in 6.9.5 Calculation results for Ground station receivers) and proposed signal

manipulation path in Ground station.


Based on the model of path between space station and ground station and parameters of

both stations, we calculated losses on the path between the communicating entities.

Based on losses and parameters of both stations, we are able to predict emitting powers on

both stations that are necessary for proper operation of the link.

Because of the fact that exact orbit that will be used by the satellite is not yet known, we

did all calculations for the worst-case scenario, therefore for the farthest orbit possible. In

better case, less attenuation will be necessary at the ground station.

The distance between ground station and space station was during the calculations

expected to be, depending on the orbit used, between 300 and 1200 km (later between 600

and 1000 km, which adds about 0.5 to 0.6 dB of expected path losses to initial calculations

done for 300 to 1200 km orbit). The communication will be happening in VHF, UHF and

S-band frequency ranges.

6.9 GROUND STATION RECEIVER

Attenuation in the receiving system is selected according to the signal level at the vicinity

of ground station, and expected signal level at the SDR, which is must be larger than -31.5

dBm [25].

6.9.1 RECEIVING SYSTEM ATTENUATION

The calculation of receiving system attenuation levels and noises was done based on the

available data about components. Each component has certain attenuation (either positive

or negative) in frequency range where our signals will pass the components.

We used predicted noise level and predicted signal level as the input to the system. Then

we attenuated both signals in according to the data by summing the attenuation levels of all

components on the path. For LNA, we also have Noise Figure information [29], which tells

us how much the SNR is lowered at the output of LNA. The result was sum attenuated


signal level, attenuated noise level, and sum of noise figures, calculated as (8.18):

S L=S input+G1+G2+...+L1+L2+...N L=N input+G1+G2+...+L1+L2+...

N F=N F1+N F2+...

(8.18)

Where:

• SL is signal level, NL is noise level and NF is noise figure

• Gx is gain at the component on the signal path in the receiver (antenna, LNA, …)

• Lx is loss at the component on the signal path in the receiver (cable, switch, …)

• Sinput is signal input, based on satellite output power and losses on the signal path

between transmitting and receiving antennas.

• Where Ninput is noise input, calculated as described in 6.9.2 Received noise at the

antenna.

From this, we can calculate final SNR in dB as (8.19):

SNR[dB]=S L [dBm]

N L [dBm]

−N F [dB]

(8.19)

The results of calculations for both UHF and S-band reception of the signals in ground

station can be found among appendixes, in A.1 Ground station part – downlink.

6.9.2 RECEIVED NOISE AT THE ANTENNA

Based on [27], noise levels on the surface of Earth are usually around -130 dBm to -140

dBm at bandwidth around 10 kHz, with higher values at higher bandwidths. The best way

to figure out noise level is to measure it, but unfortunately, this requires highly sensitive

equipment we don't have so we've had to use estimates.


We used value of -130 dBm of noise for UHF band, and -126 dBm for S-band based on

[27], page 19 (urban area).

Anyway, the more pointed the antenna is, therefore the more gain it has in particular

direction and the less gain it has in other direction, the lesser is the angle in which

unwanted (noise) signals will enter the antenna system. This is particularly obvious in

pointed S-band communication system with narrow and high-gain antenna on the ground

station.

Therefore we calculated the noise levels entering antenna as follows:

N entering=N present⋅φ

360°(8.20)

where:

• Nentering is noise that enters antenna and is then attenuated at the antenna with

antenna's gain,

• Npresent is present noise level at given frequency in given bandwidth, in our case -126

dBm or -130 dBm,

• Φ is beamwidth.

6.9.3 SYSTEM NOISE

Some noise also occurs in the receiving system itself, in both ground station and space

station. Also the amount of this noise, called system noise, is dependent on the bandwidth

of the receiving signals, and system temperature.

We can calculate noise level from temperature using the equation [28] (8.21):


(8.21)

where:

• P is the power in watts,

• B is the total bandwidth in Hz, over which noise power is measured,

• kB is the Boltzmann constant in joules/kelvin,

• and T is the noise temperature.

We found out that receiving system noise finally doesn't have big implication on SNR at

the SDR module, even for 50 MHz bandwith signals in S-band. System temperature was

calculated in the following way (8.22):

T a⋅α+T 0⋅α+T LNA+T ComRcvr

GLNA

10Lcable

10

(8.22)

where:

• Ta is antenna or sky temperature, calculated from measured noise level and

bandwidth, in our case estimated to 70K

• α is is transmission line coefficient, based on transmission line losses which are in

our case around 2 dB, therefore coefficient is 0,6331.

• T0 is ground station feedline temperature, in our case 290K because system will

work at around 20 degrees C. During the summer, the temperature will be higher.

• TLNA is LNA temperature, which is also 290K in our case.

• TCommRcvr is Communication receiver Front End temperature, which is also 290K in

our case,


• GLNA is LNA gain, which is in our case combined gain of LNA and downconverter,

and sums to 40 dB.

• Lcable is the loss of waveguide on the path from LNA to SDR input, in our case 11

dB.

We find out that System Noise Temperature is 441 K, from which we can calculate system

noise level of around -65 dBm. This noise level is finally insignificant compared to

attenuated noise level from outside sources, which based on our calculations sums to

around -40 dBm at the receiver input for both UHF and S-band receiving systems.

6.9.4 SCHEME OF GROUND STATIONS RECEIVERS AND TRANSMITTERS ANALOGUE SIGNALS PATH.

Scheme is shown on Figure 8.3. Calculations can be found in Appendixes.

Figure 8.3: Analogue signals path in Ground station.


6.9.5 CALCULATION RESULTS FOR GROUND STATION RECEIVERS

To achieve signal level of around -100 dBm behind receiving antennas on the Ground

station, satellite's EIRP needs to be:

300 mW on UHF,

800 mW on S-band.

On the downlink, we need additional attenuation of signals of 70 dB in UHF band and

additional attenuation of 23 dB on S-band.

The noise in the receiving system sums to around -40 dBm at the receiver input, while

signal at the receiver input is higher than -30 dB on all branches, so wanted SNR at the

receiver is achieved.

With the expected levels of noise in the proximity of receiving antennas of the Ground

station, with characteristics of antennas taken into account, SNR of the signal that will

reach antenna in UHF band is 18,3 dB and SNR that will reach S-band antenna is 13,8 dB.

Final noise figure for UHF band is 16,4 and for S-band is 11,7 dB, which exceeds the

receiver's requirement of Noise figure of 5 to 7 dB.

Further calculations can be found in appendixes.

6.10 GROUND STATION TRANSMITTERS

Based on the path calculations and satellite receiving system, we've had to achieve output

signal power of 66 dBm at VHF and 80 dBm at S-band. Output from SDR has power level

of 18 dBm, so proper power amplifiers were selected and path gains and losses were

calculated in similar way as for receiving systems (8.23):

S L=S input+G1+G2+...+L1+L2+... (8.23)


where:

• SL is signal level

• Gx is gain at the component on the signal path in the receiver (antenna, LNA, …)

• Lx is loss at the component on the signal path in the receiver (cable, switch, …)

6.10.1 CALCULATION RESULTS FOR GROUND STATION TRANSMITTERSFor proper function of the link, the necessary powers at the Ground station output

(antenna), to achieve -100 dBm signal level on the satellite, according to the path loss, are:

66 dBm on VHF band, and

80 dBm on S-band.

Further calculations can be found in appendixes.


7 MODULATION

7.1 WHY DO WE NEED MODULATION

In electronics, modulating means changing the properties of some carrier signal with some

other information signal. We can change amplitude, phase and frequency to carrier signal

in order to convey information.

The carrier signal can generally be written as (9.1):

(9.1)

where A(t) represents amplitude that's changing with time, and θ(t) represents a changing

angle. The angle can be written as a combination of frequency and phase as shown on

(9.2):

(9.2)

Regarding the information signal we are transferring, we can distinguish between analogue

and digital modulations. Today, most of usage is in the field of digital modulations.

With digital modulations, we want to convert the information signal into the form, which is

the most suitable for transfer over a certain communication channel. We can transfer some

signal in its baseband, or in higher frequency bands. In baseband, only impulse shaping is

important, while in higher frequency bands, it is necessary for those impulses to be

modulated with carrier sinus signal. We can achieve multiple advantages by modulating


signals, like for example adaptation of the signal to the characteristics of the

communications channel or frequency multiplexing [30].

7.2 WHICH MODULATION TO CHOOSE

The basic criteria for comparasion of different modulation techniques are:

• spectral effiiciency η can be defined as the number of bits per second, that can be

transferred in a selected frequency bandwidth. It tells us, how efficiently some

modulation uses the bandwidth [30]:

η=transfer speedbandwidth [bit/s/Hz] (9.3)

Spetral efficiency is dependent on definition of seleted band definition.

• power efficiency tells us how much power a system requires to transfer

information at certain speed in environment with some level of noise. It is defined

as SNR level, needed to achieve certain bit error level when transferring signal over

a Gaussian channel2. [30]

Besides spectral and power efficiency, there is a couple of other criteria to consider when

choosing modulation:

• sensitivity to Doppler frequency shift and Rayleigh field fading

• sensitivity to selective field fading

• system complexity

• sysem power usage

• etc ...

2 For BPSK, further explanation can be found in chapter 12 Bit Error Rate for BPSK

modulation on page 68.


7.3 SYMBOLS, BITS AND BAUDS

Symbol and bit can both be represented by sinusoidal or wave functions. However, the

difference between the two is that the symbol represents a unit of transmission energy and

the bit represents a unit of information. Symbol therefore means energy on the medium,

which represents bit(s) of information.

The analog signal shape in communications therefore stands for a certain number of bits

and is called a symbol. Such a symbol can represent one bit, or multiple bits of

information. In BPSK, one symbol represents one bit.

A baud is the symbol rate of a communication system in symbols per second. For example,

sending 50 symbols per second equals 50 bauds.

To sum up: modulation is all about packing bits into symbols, so the symbols can be

transported over the medium [31].


8 BPSK MODULATION

8.1 PSK – PHASE-SHIFT KEYING

Phase-shift keying is a digital modulation scheme which uses changes in the phase of the

carrier wave in order to transfer data. The carrier wave is usually a signal of higher

frequency than the data which is being presented by modulation.

PSK is a digital modulation scheme. That means that it uses a finite number of distinct

signals to represent digital data. Each signal is mapped into unique pattern of binary digits.

In PSK, there are more (at least two) phases used, where each phase usually encodes an

equal number of bits. One pattern of bits, which is in PSK presented by a particular phase,

forms a symbol [30].

8.1.1 MODULATOR AND DEMODULATOR RELATIONSHIP

The modulator and demodulator should therefore have the same (mirrored) mappings of

symbol to phase (at modulator) and phase to symbol (at demodulator).

8.2 BPSK – BINARY PSK

The simpliest form of PSK is Binary phase-shift keying (sometimes called Phase Reversal

Keying or 2PSK). This modulation has two constellation points and is also the most robust

PSK modulation. This means that the highest level of noise or distortion must be present to


make the demodulator reach an incorrect decision. It uses two phases, separated by 180°. It

doesn't matter exactly where the constellation points are positioned, but on Figure 10.1

they are positioned on the real axis at 0° and 180°. The bad thing about this modulation is,

that it is only able to modulate 1 bit per symbol and is as such unsuitable for high data-rate

applications.

The data signal is therefore transferred using two different phases, which can be described

as cosine signal with different leading sign [30], as with (10.1):

(10.1)

In time-domain, the carrier signal and baseband (modulation) signal, as well as their

product, which presents a BPSK signal, is shown on Figure 10.2.

Figure 10.1: BPSK constellation.


Because of the fact that communications channel introduces an arbitrary phase-shift to the

signal, the demodulator can be unable to tell which constellation point presents which

symbol. That is the reason why the data is often differentially encoded prior to modulation.

BPSK provides the lowest probability of error for a given received signal level over one

symbol period and is as such useful also for deep-space communications [33].

We can obtain frequency spectrum of BPSK by convolution of carrier spectrum and

symbol spectrum. In our case, the carrier is a pure sinus, therefore in frequency spectrum it

is presented by an impulse, located at carrier frequency. Convolution of any spectrum with

Figure 10.2: Carrier signal, baseband modulation signal and resulting BPSK modulation. [32]


impulse centers that spectrum at the frequency of impulse. Therefore, in BPSK, we have

baseband spectrum, centered at carrier frequency.

The baseband symbol spectrum is a bit more complicated, because it consists of

rectangular pulses. One rectangular pulse - square wave - is in frequency space presented

as an infinite series of weighted impulses at all odd harmonics of the fundamental

frequency. But in BPSK case, baseband consists of rectangular pulses of widths that are

integer multiplies of one symbol. The resulting spectrum contains besides fundamental

symbol frequency and all its odd harmonics also all integer sub-harmonics of the

fundamental, along with their odd harmonics.

The spectrum is shown on Figure 10.3. The carrier frequency in the example is 1500 Hz,

while the symbol frequency is 600 Hz. The sampling frequency is 8 kHz [32].

Figure 10.3: BPSK frequency spectrum (symbol frequency: 600 Hz) [32]


The bandwidth of signal so far is therefore infinite. However, in practice this is

unacceptable because of the fact that for communication, only a finite amount of

bandwidth is available. Therefore the baseband signal must be filtered. This changes the

shape of the pulses into a form which is has practically more acceptable spectral properties

than a rectangular pulse [32].

8.3 DIFFERENTIAL CODING

Because of the fact that the constellation at the receiver can rotate because of some effect

in the communications channel, it is useful to use differential coding to avoid impact of

this effect.

Regarding the description of coherent PSK demodulation system, the system compares the

received signal to the reference carrier signal in terms of phase. That means that if the

carrier signal is recovered incorrectly, the result can be an inverted received data.

By using differential coding, the receiver depends only on the difference between current

phase and previous phase, and not on the exact values of the phase. For example, in

differentially-encoded BPSK, a binary '1' may be transmitted by changing phase for 180°

and a binary '0' may be transmitted by not changing phase (therefore changing phase for

0°). That way, the receiver is immune to the constellation rotations that occur in the

communications channel.


9 NRZ ENCODING

Non-return-to-zero encoding is simple binary line encoding, which means that it is used to

encode data in baseband. It uses two voltage levels for representation of two values: '0' and

'1'. NRZ encoded bit stream is shown on Figure 11.1.

In NRZ, therefore, only two states are used, with no third, neutral or rest condition.

Therefore, the NRZ uses more energy than RZ (return-to-zero) encoding and is not self-

synchronizing. This means that some additional synchronization technique is required to

avoid bit slip (which is caused by clock variations of the transmitting and recieving

devices).

For example, in case of a long sequence of the same value, it is impossible to synchronize

because the value at the recieving end doesn't change much.

Figure 11.1: NRZ encoding example [34]


The NRZ code requires, however, only half of the bandwidth, required by the Manchester

code. NRZ-Level as an encoding can be used either in synchronous or asynchronous

transmission environment, therefore with or without explicit clock signal present. It is

therefore not so important how the encoding acts on the clock edge, but rather what state is

being sampled at the time of sampling on the recieving end [34].


10 OVERVIEW OF TASKS AT THE RECEIVER

At the receiver, the signal must go through several stages of adjustments and processing

before the actual demodulation occurs. The stages are the following:

1. Automatic gain control scales signal to a desired power level. This is usually done

in analog domain, because AD converters have a limited dynamic range and the

input signal must be within that range. By adjusting gain, clipping (because of too

high signal strenght) and distortion (because of too low signal strength) are

avoided.

2. Timing recovery is the stage in which symbol synchronization is obtained. In order

to achieve synchronization, the sampling frequency and sampling phase must be

correctly adjusted, so the sampling happens as close to the center of each symbol

period.

3. Carrier recovery is necessary because the actual signal frequency at the receiver in

practice is different than the frequency at the internal oscilator of the receiver. At

the passband, the received signal is multiplied by that receiver internal oscilator

signal and if the frequencies are not the same, the signal is near baseband with

some frequency offset. That is why the signal constellation rotates.

4. Channel equalization is needed when signal is transmitted through a multipath

channel. The received signal consists of several delayed and scaled versions of

transmitted signal. We can look at the channel as a linear filter and the equalizer

plays the role of an adaptive filter, which attempts to remove intersymbol

interference by filtering the received signal with the goal to undo effects of the

multipath channel.


11 DEMODULATOR IMPLEMENTATIONS

Two types of systems for different demodulation schemes exist in general:

• coherent: the receiver compares the phase of the received signal to a reference

signal. Regarding this comparison, the demodulator maps the phase difference to

the symbol which is represented by that phase difference. However, the coherent

demodulator requires constant reference wave.

• differential PSK: this system considers changes in phase of a single broadcast

waveform as the significant items. The demodulator therefore determines the

changes in the phase of the received signal, rather than the phase relative to the

reference wave. That's why this scheme is called „differential“ PSK. Due to the

fact, that there is no need for demodulator to have a copy of the reference signal to

do its job, differential PSK system can be significantly simpler to implement. On

the other hand, it does not determine the exact phase of the received signal and

produces more erroneous demodulation results.

11.1 COHERENT DEMODULATOR IMPLEMENTATION

The coherent demodulator is shown on Figure 13.1. The demodulator does the

multiplication of the modulated input signal s(t) with the reference signal cos(ωt).

If the signal s(t) in some moment is in phase with reference cos(ωt), then the multiplier

multiplies together two positive or two negative values, which results in a positive value.

If the received modulated signal s(t) in some moment differs in phase from reference signal


cos(ωt) for 180°, the multiplier multiplies together negative and positive value, which

results in a negative value.

The output from multiplier i(t) is then integrated over a time period of one bit. This results

in an analogous value, which presents an input to threshold detector. The output of treshold

detector si(t) presents received bit value.

If NRZ encoding is used, then value of „0“ is the threshold value for distinguishing bits 0

and 1 [35].

11.2 CARRIER RECOVERY FOR COHERENT DEMODULATION

In an ideal communications system, the transmitter and receiver would have perfectly

matched local oscillators in frequency and phase, to perform coherent demodulation. In

practice, however, this is not the case, because they rarely share the same carrier frequency

oscillator. The fact is that the receiving systems are usually independent from transmitting

systems. Also there are effects like Doppler shift on the path of the signal, sent by a

moving transmitter, that contribute to frequency differences in radio frequency

communications systems.

In coherent demodulation, we need recovered carrier wave in order to accomplish

demodulation process.

In some cases, the carrier signal information is sent synchronously with the data

information on separate line, like this is the case in some serial telecommunication links. In

some other cases, we need to recover the carrier from the received modulated data signal.

For carrier recovery, we need a carrier recovery system - a circuit which estimates and

Figure 13.1: Coherent demodulator


compensates the frequency and phase differences between a carrier wave of the received

signal and signal, created by receiver's local oscillator.

Basically two kinds of carrier recovery methods are known:

• non-data-aided methods which are typically used for simple carrier recovery

schemes or for initial coarse carrier frequency recovery, and

• decision-directed methods: in this kind of carrier recovery methods, the output of

symbol decoder is fed to a comparison circuit which outputs the phase difference or

error between the decoded symbol and the received signal. This difference is then

used to calibrate local oscillator [36].

11.2.1 PHASE-LOCKED LOOP

Phase-Locked loop (PLL) is a feedback phase and frequency control loop, which generates

a signal whose phase is related to the phase of input signal. PLL has many uses in

communications, for example in frequency demodulation, multiplication and division,

syntetization and synchronization. It can be used to generate a frequency that is multiple of

input frequency, or track an input frequency.

PLL consists of the following main components:

• some kind of an error detector (phase detector),

• a loop filter

• a VCO (Voltage Controlled Oscillator)

PLL can be a fully analog, partially analog partially digital, or fully digital circuit [37]. It

operates in the following way. Simple model is shown on Figure 13.2. The error detector

generates signal whose value represents and estimate of the phase error (phase difference

between the phase of incoming signal and the signal, generated by VCO). This phase error


estimate is then filtered and fed back to the VCO in order to reduce the phase difference in

the next cycle [38].

PLL modelIf we refer to Figure 13.2, we can say that an input signal is presented as (13.1):

(13.1)

where ωc presents radian frequency of the carrier and θ presents its arbitrary phase. This

input signal is applied to the receiver input and is multiplied by the local reference, which

is (13.2):

(13.2)

The output of PD, which on Figure 13.2 represents an „error detector“, is (13.3):

(13.3)

The double frequency term in practice leads to a ripple in the error control signal and is in

Figure 13.2: Simple PLL model. [37]


practice usually compensated with a LPF (lowpass filter). The loop filter is also a LPF with

very narrow bandwidth. If we assume an ideal filter and neglect the double frequency term,

we get (13.4):

(13.4)

The loop filter is a lowpass filter for error signal. Its design, order and related parameters

are very important for wanted PLL design [38].

Second-order PLLThe order of PLL is defined as the number of „perfect integrators“, n, in the loop. An

integrator defined by 1/s is a pole, placed at s = 0, or direct current (DC), and is

represented by VCO. The n of the PLL is therefore one more than the order of the loop

filter, so the n-th order PLL may be constructed as (n-1)th-order loop filter. If we use a

first-order RC filter as a loop filter, the result is a very well known and widely

implemented second-order PLL. This kind of a loop is useful for wide range of

applications, including carrier recovery. The simplest, first-order PLL, may be constructed

with a loop filter, equal to constant, but it is not useful for carrier recovery because it can't

compensate frequency ramp and poorly compensates frequency offset.

Third-order PLLThe third-order PLL is not used as commonly as second-order PLL, but has some

advantages. For carrier recovery, the important advantage is its capability of tracking

frequency ramp with zero error in steady-state, while second-order PLL can do this only

with a finite steady-state error. The loop filter, used in third-order PLL, is a second-order

filter.


Digital Phase-Locked Loops

The general DPLL block diagram is shown on Figure 13.3. With the exception of the

sampler, DPLL is similar in form and operation to general PLL. However, the analog

elements behind the sampler are replaced with digital elements, which provides several

benefits: elimination of problems because of component aging and temperature drifts, more

flexibility in design and testing and greater system integration using VLSI (Very Large

Scale Integration) techniques [38].

Several different types of DPLLs exist because of different implementation of sampling

processes. In , four categories of DPLLs are presented:

• Nyquist rate DPLL (NR-DPLL): evolved from analog PLLs, where a sampler is

placed before the loop and samples the signal at uniform Nyquist rate.

• Flip-Flop DLL (FF-DPLL): first method, depicted in Figure 13.4, that employs a

non-uniform sampling method and doesn't need a conventional phase detector; the

phase error is derived from the duration between the Set and Reset times of Flip-

Flop, triggered by reference and input signal while counter is triggered by internal

clock.

• Lead/Lag DPLL (LL-DPLL): at every cycle, the phase detector decides wether

the input signal is leading or lagging the reference signal. Integrate-and-dump

Figure 13.3: Simplified DPLL block diagram. [38]


circuit is used for this, followed by a threshold detector. Its block diagram is

depicted in Figure 13.5.

• Zero Crossing DPLL (ZC-DPLL): tracks positive zero crossings of the received

signal. Sampler is used as the phase detector. NCO (Numerically Controlled

Oscillator), which replaces a VCO, derives the sampling instants. Its block diagram

is depicted in Figure 13.6.

Figure 13.4: FF-DPLL block diagram [38].

Figure 13.5: LL-DPLL block diagram [38].


Digital Tanlock LoopWith high-speed digital logic circuits, one more noteworthy carrier recovery structure, the

Digital Tanlock Loop (DTL), became feasible. DTL is the structure that relies on explicit

arctangent operation to produce phase error term and was shown ([39]) to have several

advantages over traditional closed-loop carrier recovery circuits, like operating without

separate Automatic Gain Control (AGC) due to the fact that arctangent operation is used

[37]. Digital tanlock loop is shown on Figure 13.7.

Figure 13.6: ZC-DPLL block diagram [38].

Figure 13.7: Digital tanlock loop [48].


11.2.2 NON-DATA-AIDED METHODS

Multiply-filter-divideThis is an example of open-loop carrier recovery, which means that it uses only the current

state of a system (and not also previous states) to get the job done. This is good for burst

transactions, because of generally lower acquisition time compared to close-loop

synchronizers [36].

The method consists of the following steps [36]:

1. applying a non-linear operation to the modulated signal, to create harmonics of the

carrier frequency with modulation removed

2. filtering carrier harmonic using bandpass filter

3. dividing filtered carrier harmonic to recover the carrier frequency

Costas loopA Costas loop can be used for carrier phase and frequency recovery, as well as

demodulation. It is somewhat similar to PLL that uses coherent quadrature signals to

measure phase error. The PLL is not useful for PSK signals because of constant changing

of phase. Costas loop removes modulation and provides recovered carrier PSK signal. The

measured phase error is used to correct the loop's oscillator. Once the quadrature signals

are properly aligned and recovered, the Costas loop successfully demodulates the signal

[36].

Figure 13.8 depicts a basic Costas loop. In [40], implementation of Costas loop in sample-

based technology is described. Usage of two identical filters is harder to implement with

discrete filters, but in sample-based approach, matching the two is quite simple.


The flow of the signal through the loop is depicted using the circles with numbers.

Circle 1 shows the input signal entering the Costas loop. In the example, this is a simple

sine wave.

Circles 2 and 3 show the input signal, multiplied with cosine and sine waves from the

NCO (Numerically Controlled Oscillator) and the resulting equations after multiplication.

The system sample rate should be at least twice of the input frequency (Nyquist).

Circles 4 and 5 show signal after it traversed the LPF (Low Pass Filter). The resulting

expression shows that there is only low-frequency component left, because it assumes that

the filter is ideal. In practice, it's not ideal, but we assume that higher-frequency

components are sufficiently suppressed.

Circle 6 shows the mixed signal from both arms. Regarding the resulting expression on

Figure 13.8 and the fact that sin(0) = 0, the resulting signal error is a function of the

difference between angles θ and ϕ.

This error is then employed by NCO in order to push the loop in the direction, opposite to

the error between internally generated and received signal, and therefore bringing the loop

Figure 13.8: Costas loop [40].


to lock.

Squaring loopThe squaring loop does theoretically have the same noise performance as the Costas loop,

but is a bit more complex. The signal is first bandpass filtered to reduce noise, and then

squared. This produces a discrete spectral line at 2ωc which is then filtered with another

bandpass filter. The second filter suppresses frequencies outside of the range in which 2ωc

spectral line may occur. Output of second bandpass filter is then applied to PLL, which is

tuned to 2ωc and yields a phase estimate of 2θ. The output of PLL is then divided by 2 to

produce the recovered carrier.

Because of the multiplication and division operations, there is a 180° phase ambiguity [37].

Squaring loop is shown on Figure 13.9.

Open-loop carrier recovery structuresSeveral open-loop carrier recovery structures exist, and based on [41], we will describe

proposals, based on the maximum likehood (ML) principle. The described estimators are

based on [37] named as first-order, modified first-order and second-order ML Phase

Figure 13.9: Squaring loop structure [37].


Estimators, because they have precise performance analogies to first-order, modified first-

order and second-order PLL-s and are drawn in [37]. The estimation structures depend on

whether low SNR or high SNR conditions are assumed.

First-Order ML Phase Estimator, applied to binary phase modulation, is shown in Figure

13.10. It includes an averager which computes a sum of squared inputs over N symbol

observations. The squaring operation doubles the phase of the input sample, which is later

compensated by square root operation. The squaring and rooting process produces 180°

phase ambiguity, similar to Costas and squaring loops.

Because in reality, carrier phase θ is usually not constant for the length of time over which

N observations of rk are made, the First-Order ML Phase Estimator can not track frequency

step or phase change without error when in steady-state, similar to first-order PLL. The

modification to ML phase estimation structure is necessary to make it suitable for real-

world conditions, which leads us to Modified first-order ML phase estimator as it's

denoted in [37] that is suitable for small residual frequency offset and is shown in Figure

13.11 and includes additional 180° phase-jump detector.

The third proposed structure is in [37] denoted as Second-order ML phase estimator and is

suitable for larger frequency offset in received carrier signal during steady-state operation.

Besides the components, included in Modified first-order ML phase estimator, this

structure includes also open-loop frequency estimation scheme, which is added in parallel

and attempts to estimate 2Δθ. The structure is shown on Figure 13.12.


Figure 13.10: First-order ML phase estimator [37].

Figure 13.11: Modified First-order ML phase estimator [37].


In [37], performance analogies with PLL-s are shown, as well as the fact is presented that

ML Phase Estimator structures provide ML estimates of signal phase for binary phase

modulation in AWGN channel.

Phase ambiguityFor purely coherent detection of phase modulated signals with locally generated

references, the shortcoming which may lead to problems at carrier recovery is 180° phase

ambiguity. It may be handled for example by using training sequence or some other

method, but is generally a problem when Costas loop or squaring loop is used. [37]

Figure 13.12: Second-order ML phase estimator [37].


11.3 DIFFERENTIAL PSKBy using differential PSK, we can avoid to the problem of carrier recovery by using

differential encoding. This means that absolute phase is not important. Before we modulate

digital signal, we encode it in a way that every bit value depends on the value of current

and previous bit. This can be achieved by using XNOR (Exclusive-NOR) gates, which

output '1' if the input bits are different and '0' if the input bits are the same. The

demodulator is realized as a comparator between the phase of current and previous symbol.

Differential PSK demodulation is much simpler than coherent, because we don't need

carrier recovery. The bad thing is, that it is not so resistent to noise as non-differential PSK.

Figure 13.13 Shows the differential PSK processing systems.

There is a proposal for implementation of BPSK demodulator with digital technology,

described in [42]. The described demodulator block diagram is essentially the same as the

generally known BPSK demodulator, and is shown on Figure 13.14.

Figure 13.13: Differential PSK modulator and demodulator [40].

Figure 13.14: Differential BPSK demodulator [42].


However, the Delay of one bit is realized in a way which uses digital, sampled data.

Because of the fact, that Delay of one bit must be strictly synchronized with the input, the

differential BPSK demodulation is difficult in practice. If the input carrier is (13.5):

(13.5)

then the delayed signal must be shifted in phase by θ, therefore the delayed signal must be

(13.6):

(13.6)

Based on that, the output from the multiplier is the following:

(13.7)

Because ideally, θ is 0° or 0°, we can ignore the second term. The first term comprises of

one DC component, and one large ripple.

The signal from multiplier is then used for data clock recovery circuit. However, the large

ripple causes problems with the stability of clock recovery hardware.

In the proposed method, the delay is achieved by using ADC (Analogue to Digital

Converter) and ELPD (Electrically Programmable Logical Device) Delay Line, which is

17 bit long and 8 bit wide. It is shown in Figure 13.15.


The sampling frequency at ADC must be chosen in a way, that one sampling period delay

corresponds to 90° phase shift of the carrier wave. If there is a sine waveform present at

taping point 1 in ELPD Delay Line, then a cosine waveform is present at taping point 2.

The signals at taping points 1 and 16 will therefore be In-Phase (I) components, while

those at taping points 2 and 17 will be Quadrature (Q) components.

The following waveform is present at the adder in Figure 13.15:

(13.8)

where θ presents phase shift of carrier over 1 bit period of time and ϕ presents variation

from the 90° phase shift. In practice, θ will be either 0° or 180° for BPSK data, and ϕ will

be zero. The resulting waveform at the adder will therefore be (13.9):

(13.9)

That way, the ripple at 2ωc is removed completely. This waveform is then used by PLL for

carrier recovery, which is necessary for correct clocking of digital components (starting

with ADC). It is also integrated and dumped after one bit period, and on the output of

Integrate and Dump component, there is demodulated data. [42]

Figure 13.15: BPSK demodulator wit EPLD Delay Line [42].


12 BIT ERROR RATE FOR BPSK MODULATION

In this section, I will explore the probability for errors to happen at the recieving

(demodulating) end of BPSK communication in communications channel with Additive

White Gaussian Noise (AWGN). AWGN stands for noise that gets added to the signal that

represents data in our case (that's why „Additive“), that has flat spectrum for all

frequencies (that's why „White“), and that the values of noise follow the Gaussian

probability distribution (that's why „Gaussian“).

In probability theory, the Gaussian is also called a normal distribution and represents a

continuous probability distribution with bell-shaped probability density function, called

Gaussian function, which is:

(14.1)

Parameter μ represents the mean value, which presents the location of the peak of Gaussian

function. Parameter σ2 represents standard deviation. In our case, it is derived from N0,

which somehow represents the power of noise. That is because the larger the N0, the larger

the σ2 is and the greater the probability for larger power of noise is. In standard normal

Gaussian distribution, σ2 = 1.

Bit Error Rate is the number of bit errors that happen per transferred amount of bits in

some amount of time. The more noisy the channel is and the weaker the transmitted signal

is at the recieving end (therefore the lower Signal to Noise Ratio - SNR), the greater


probability generally is that there will be errors on the receiving end. What follows is a

mathematical explanation of how the AWGN affects BER at the receiving BPSK end.

Considering Figure 13.16, we can say that the binary digits 0 and 1, modulated using

BPSK, can be represented by analog levels +√Eb and -√Eb. In the channel, the noise n gets

added to the transmitted signal, resulting in the signal y. Noise n is AWGN in our case. The

demodulator then demodulates the noisy signal y and outputs the result.

We can say that the received signal is y = s1 + n when bit 1 is transmitted, and y = s0 + n

when bit 0 is transmitted. In Section 5.2.1 of [43], the derivation is provided for

computation of probability of error. The conditional probability distribution functions

(PDF) of y for the given cases are the following:

(14.2)

If s0 and s1 are equaly probable, optimal decision treshold is at value 0. Therefore if the

received signal is less or equal to 0, then the receiver assumes that s0 was transmitted

(y<=0 => s0). If the received signal is greater than 0, then the receiver assumes s1 was

Figure 13.16: Noise in the channel.


transmitted (y>0 => s1).

On Figure 13.17, the upper part of illustration shows the boundary between symbols at

value 0 and the symbols on the x axis.

The lower part of illustration shows:

• p(r/s0) – the probability of recieving signal s0 with certain value of level -√Eb and

• p(r/s1) – the probability of recieving signal s1 with certain value of level +√Eb

• Blue area: probability that s1 was transmitted and there was an erroneous

demodulation

• Green area: probability that s0 was transmitted and there was an erroneous

demodulation

The probability of error, given that s1 was transmitted, is shown in the blue region of

Figure 13.17 and is based on the previously described conditional probability function and

Figure 13.17: Treshold, probability of received signals and error probability [44].


can be expressed as the (14.3) [44]:

(14.3)

We have the similar situation when computing the probability of error, given that s0 was

transmitted. It is shown in the green region of Figure 13.17 and can be expressed as (14.4):

(14.5)

The complementary error function erfc(x) is the following (14.5):

(14.5)

Total probability of bit error can be computed by multiplying the probability of sending

one of the symbols with probability of error when that symbol is sent, and adding those

values together for both symbols:

Pb= p(s1)⋅p(e∣s1)+ p(s0)⋅p(e∣s0) (14.6)

[44]

If we assume that s0 and s1 are equally probable, we can say that p(s0) = p(s1) = ½, we can


express bit error probability as (14.7):

(14.7)


13 TELEMETRY SYNCHRONIZATION AND CHANNEL CODING

Telemetry from satellite to ground station will be transfered in accordance to recommended

standard CCSDS 131.0-B-2 [45].

Inside FPGA, the final product of thesis, demodulator, will also perform some of Data Link

Layer functions from ISO/OSI reference model, namely telemetry (TM) synchronization

and channel decoding, as shown on Figure 15.1. Those functions are provided in TM

Synchronization and channel coding layer of CCSDS protocol stack. Demodulator will

work in accordance to standard CCSDS 131.0-B-2.

The following functions are covered by TM synchronizadion and channel coding sublayer

[45]:

• error-control coding

• frame validation

• synchronization

• pseudo-randomizing


13.1 ERROR-CONTROL CODING

The standard recommends four types of error-control coding: Convolutional coding, Reed-

Solomon coding, Turbo coding and Low-Density Parity-Check (LDPC) coding. Which

coding is used for particular task depends on performance requirements, like the capability

of indicating uncorrectable errors (Reed-Solmon and high rate LDPC codes), minimal

bandwidth expansion because of coding (punctured convolutional, Reed-Solomon and high

rate LDPC codes) and larger coding gain (which can be achieved using concatenation of

convolutional code as inner code and Reed-Solomon as outer code, or where possible

Turbo codes or LDPC codes) [45].

13.2 FRAME VALIDATION

Figure 15.1: CCSDS protocols, ISO/OSI model and our FPGA-based demodulator functions. Based on Figure2-1 in [45].


The upper layers at the receiving end need to know whether Transfer Frame after decoding

is a valid data unit or not. The Reed-Solomon and LDPC decoders can determine whether

they can correctly decode frame with very high probability, and can therefore be used also

for frame validation. If the other codes are used, the Frame Error Control Field is used

[45].

13.3 SYNCHRONIZATION

Frame synchronization is necessary for subsequent processing of Transfer Frames, as well

as for proper decoding of some codes: Reed-Solomon, Turbo and LDPC. It is also used for

synchronization of pseudo-random generator. For some coding systems, the Attached Sync

Marker (ASM) can be acqured in the channel symbol domain, therefore before decoding,

where for some ASM can be acquired in the domain of bits decoded by decoder [45].

13.4 PSEUDO-RANDOMIZING

For the proper operation of receiver system, it is important that the receiving signal has

sufficient bit transition density. This allow proper synchronization of the decoder.

Receiving signal must also be free of significant spectral lines and free of certain patterns.

The data stream at the receiver must therefore be sufficiently random to achieve proper

receiver operation. That's why pseudo-randomizer is necessary, unless it is verified that it is

not necessary in some system [45].

13.5 ORGANIZATION OF SYNCHRONIZATION AND CHANNEL CODING SUBLAYER OF THE RECEIVER

Based on [45], this Sublayer involves functions, presented on Figure 15.2. The input into

Sublayer at the recieving end is continuous and contiguous (therefore without any pauses


between subsequent symbols) stream of channel symbols from the Physical Layer. The

Sublayer perform some of the functions, depending on the mission, and delivers Transfer

Frames to the Data Link Protocol Sublayer [45].

Figure 15.2: Synchronization and Channel Coding sublayer functions [45].


14 SYSTEM SPECIFICATIONS

14.1 SIGNALS SPECIFICATIONS

• Carrier frequency: 2425 MHz

• Channel filter: 2400 to 2450 MHz

• Information signal rate: 8 kbit/s

Information signal frequency of changing: 8 kHz

The analogue path of the continuous signal through the space is simulated by using

Simulink's components, sampled with higher frequency.

• Frequency of path sampling: 5200 MHz

14.2 DOWNCONVERSION SPECIFICATIONS

Downconversion is achieved using multiplication of input signal with the signal of lower

frequency, and then filtering.

• Frequency of signal, used for multiplication: 2424.9 MHz

Filter after the multiplication:

• FIR filter with Hann window

• Cutoff frequency: 200 kHz


The signal is downconverted to 520 kHz sampling frequency.

The result:

• Carrier signal at 0.1 MHz = 100 kHz

• Data rate of the signal: 8 kHz

This signal enters the demodulator.

Arm filters:

• Order: 48 (coefficients)

• FIR Constrained Equiripple

• Passband edge frequency: 8000 Hz

• Stop magnitude: -50 dB

Filter before VCO:

• Order: 48 (coefficients)

• FIR Constrained Equiripple

• Passband edge frequency: 4000 Hz

• Stop magnitude: -50 dB

The method, used for carrier recovery, will be Costas loop, because of its relative

simplicity and the fact that principles from Costas loop for BPSK can be used to develop

Costas loop for QPSK. Implementation inside FPGA will be fully digital and will include

digital PLL as a part of Costas loop.


15 COSTAS LOOP PARAMETERS IN PRACTICE

In practice, Costas loop should be tuned according to the parameters of the input signal.

Figure 17.1 depicts digital Costas loop model, built in Matlab. Before entering the Costas

loop, the signal may be mapped to a lower frequency. By mapping the signal to lower

frequency, we may lower the sampling frequency of the Costas loop components and in

some cases simplify the digital filters used in our loop.

On Figure 17.1, there are two Discrete-Time VCOs used for generation of carrier signal.

Both are initially tuned to the expected carrier frequency of Input signal, but have different

phase: VCO 1 has initial phase of 0, while VCO 2 has initial phase of pi/2.

Filter 1 and Filter 2 are based on specification of source's baseband signal. Because BPSK

modulation only transfers two symbols, the maximal frequency of symbol alternation is the

same as data bit rate. For example, if the bit rate is 8 kbps, the frequency of the source

fSource is 8 kHz.

If we refer to frequency space representation of our baseband (modulation) signal in Figure

13.17, we see that large part of energy is present in frequency range between -fSource and

+fSource, where the center of this energy is positioned at carrier frequency when the signal is

modulated.

Because the input signal is mixed with approximate signal from VCO, the resulting

spectrum that enters the filters contains baseband signal energy in the baseband. Based on

Figure 13.17, the purpose of filter is to filter out higher-frequency signals, therefore

leaving only baseband signal energy in the signal. This means that the Filter 1 and Filter 2

are set in a way that they pass the baseband and suppress the other frequencies.

Filter 3 is set to pass the frequencies up to 1/2 fSource. In this frequency range, the error term


is contained after multiplication in Product 3.

Figure 17.1: Digital Costas loop model, based on [46].


16 BPSK SYSTEM MODEL

The BPSK system is modeled in Simulink and is shown on Figure 18.1. It consists of:

• Bernoulli Binary Generator which generates a stream of bits, which are going to

be modulated, transferred over the channel and demodulated on the other end,

• BPSK modulator block which takes the bit stream in the baseband and modulates

it with BPSK modulation at carrier frequency for transmission over channel,

• Channel block with the purpose of simulation of noise and other phenomena that

affects signal while it's traveling through Space and Earth's atmosphere,

• Downconversion block which serves the purpose of mirroring the BPSK signal to

lower frequency than is carrier frequency, which is necessary for effective sampling

of the signal and further digital processing of the signal,

• BPSK demodulator block which implements a Costas loop design, tuned at carrier

frequency that comes from Downconversion block, for signal demodulation.

Outputs demodulated bits.

• Scope which shows the bit stream that enters the BPSK modulator block, the signal

that reaches the Downconversion block and bit stream on the exit from the BPSK

demodulator. As long as the demodulation works well, the bit stream on the exit

from the BPSK demodulator block is the same as bit stream that enters the BPSK

modulator block.


16.1 BPSK MODULATOR BLOCK

The modulator for BPSK is relatively simple and is shown on Figure 18.2. It consists of

two sine waves, Sine Wave 1 with phase offset of 0 and the other with phase offset of PI/2.

The modulator is a switch, which switches between those two signals, based on the input

bit. The switch is implemented inside BPSK modulator. Signal exiting the switch is then

bandpass filtered according to parameters, described in 14 System specifications. The

bandpass filtering is necessary, because the channel for transmission of data has a limited

bandwidth.

A Scope is also present in the block, which allows us to observe the function of BPSK

modulator.

Figure 18.1: BSPK system model in Simulink.


16.2 CHANNEL BLOCK

The channel simulation exists from a custom Matlab function, RF impairment blocks and

AWGN Channel and is shown on Figure 18.3.

For channel simulation, we are using several blocks from Simulink's Communications

Blockset under RD impairments:

• Free space Path Loss

• Phase Noise

• Phase/Frequency Offset

Figure 18.2: BPSK modulator Simulink model.

Figure 18.3: Channel simulation block.


Those blocks are taking complex numbers as an input and return complex results.

The signal that traverses our channel in the simulation is a function of time with real

values: it is a signal with certain real amplitude, which is changing through time.

Because of the fact that the signal which traverses the channel in certain moment of time in

doesn't accumulate any information about the previous or future signals, therefore because

of the fact that it has no time corelation, simulation of different frequency- or phase-

related phenomena is not directly possible.

That's why we translate the input into a complex number, based on our knowledge about

the signal which is produced at the modulator side.

Because we know, what is the maximal amplitude of the signal at the exit of the modulator,

and because we know that the signal itself is either a sine or cosine signal, we can easily

translate the real amplitude of the signal into the complex number, according to

Pythagorean theorem:

function [Re,Im]= myPitagora(amp)%#emlfullAmp = 0.6;Im = sqrt(fullAmp*fullAmp - amp*amp);Re = amp;

The Real amplitude of the signal is the amplitude in a given moment, while the Imaginary

amplitude of the signal is derived from the knowledge about full amplitude of the signal

which is exiting the modulator.

At this point, the signal consists of Real (x) and Imaginary (y) component, and an angle φ

between them, as shown on Figure 18.4. As such, it is appropriate for simulation of

different RF impairments.

After the signal is processed with RF impairment blocks, the real component of the signal

is used for further simulation, because it contains the information about the amplitude of


the signal in certain moment.

Free Space Path Loss block

At this block, the loss is expressed and defined in decibels. The reciprocal of the loss is

applied as a gain, therefore the loss of 20 dB reduces the signal by factor 10, or in the other

words, gain of 0.1 is applied to the signal.

Phase noise block

This block adds a noise with spectrum that has a 1/f slope, where f is Frequency offset,

defined in Hz. The level of noise is defined with Phase noise level field in dBc/Hz, the unit

used to define the power ratio of an information signal to the power of carrier signal.

Phase/Frequency Offset block

This block simply applies a phase offset (defined in degrees) and frequency offset (defined

in Hz) to the current signal.

AWGN Channel

Figure 18.4: Imaginary number representation.


This block adds White Gaussian Noise to the signal. To define the SNR (Signal to Noise

Ratio) of the signal, the Input signal power should be defined, referenced to 1 ohm, in

watts. Because we know the amplitude of our input signal, we know that the power of the

signal is (18.1):

P=V 2

R(18.1)

where P is power, V is voltage and R is resistance, and resistance in our case by definition

equals 1 ohm.

The block itself works on real input signal, and what it does is the addition of random

amplitudes to the input signal at each sample. The random amplitudes are leveled based on

input signal power and defined requested SNR.

16.3 DOWNCONVERSION BLOCK

The purpose of this block is mirroring the signal from original frequency to lower

frequency which is suitable for demodulation with Costas loop. This is achieved by mixing

the input signal with Sine Wave. The Sine Wave has frequency, which is the difference of

input frequency and wanted output frequency. By mixing two signals with different

frequencies f1 and f2, the new frequenies at frequency at the sum f1 + f2 and difference f1

- f2, are produced. This process is known as heterodyning. Because we are interested only

in low frequency signal (therefore the difference between input frequency and Sine Wave

frequency), the Digital Filter is used to filter out the high frequency signal. Because the

analogue path of the signal is here also modelled with digital components, which have way

higher sampling rate, the downsampling block is also included. In practice, instead of

downsampling block, there would be Analgoue-to-digital (AD) conversion. Also in this

block, the Scope is included to monitor the behaviour of the signals in the block.


16.4 BPSK DEMODULATOR BLOCK

This block implements a Costas loop design, which is described in detail on 15 Costas loop

parameters in practice. The parameters of the building blocks are based on 14 System

specifications and the Scope is included to monitor the function of the design.

16.5 MODEL IN PRACTICE

In practice, the system was simulated with speed of 8 kbps and two different channel

configurations:

• configuration without signal impairments in the channel, with the result shown in

Figure 18.6, and

• configuration with all of the described impairments present and modelled, and SNR

of 11 dB (expected SNR, based on our Link budget calculations), with the result

shown in Figure 18.7.

Figure 18.5: Downconversion block in Simulink


On the figures, the input and output logical values are shown. The input value was first

modulated to signal, which travelled over channel, was downconverted, and finally

demodulated back to represent output value of 0 or 1.

Figure 18.6: Original recovered bit sequence in system without signal impairment in the channel.


In case without signal impairments, the demodulation is working error-free. In

demodulation with signal impairments, triple detection of signal level in the output of

demodulator per bit time and averaging of detected values is proposed to create more

robust output. For higher data speeds with BPSK, higher SNR and different design of

Costas loop with different filters is required.

16.6 STEP TOWARDS IMPLEMENTATION IN HARDWARE

A step towards implementation in hardware has been made by prototyping the Costas loop

design in Xilinx System Generator. The goal was to implement the described Matlab model

with Xilinx System Generator blocks, which imposes several hardware limitations (for

example reasonable number of filter coefficients, speed, sampling and timing limitations)

on the design.

Figure 18.7: Original and recovered bit sequence with signal impairments in the channel and AWGN noise with SNR of 11 dB.


During the System Generator model design, several changes (particularly to filters) were

made to Matlab model in order to create a model which can be implemented in a real-

world, constrained FPGA.

A brief description of a model can be found in B Xilinx System Generator prototype of

demodulator.


17 CONCLUSION

In the thesis, the whole link between Ground station's computers and Space station's

computer (data handling unit) has been analyzed and ground custom station design

proposed. Multiple conclusions can follow from the analysis and proposal.

First, we have shown and calculated how to achieve high-speed pointed S-band

communications and unpointed UHF/VHF communications between Ground station and

Space station on the distance between 600 and 1000 km, where Space station has only

30x10x10 cm of volume and mass, which is lower than 4 kg. Besides, the Space station

performs multiple other functions besides communication with Ground station - therefore

it has limited resources in terms of dimensions and therefore also in terms of available

power.

Based on the results, transmitting circuit on the on the Space station should transmit with

up do 30 dBm on S-band, which is 1W, and with up to 33 dBm, which is up to 2W on UHF

band, to achieve good enough reception at the Ground station, assuming that antennas on

UHF band have 0dB gain and that S-band antennas have 6dB gain, as available.

The Ground station on the other hand will have most of the attenuation elements on both

receiving and transmitting circuit, and will need to effectively transmit with about 49 dBm

(80W) of power (EIRP) or more on S-band, and between 23 dBm and 51 dBm (from less

than 1 W, and up to 125 W) on VHF band EIRP.

High amplification will be required as a part of Ground station for all circuits, while

satellite won't have any additional elements between antenna and transceiver circuit


because of its volume limits.

Second, demodulator design is proposed to be implemented in FPGA. Because BPSK

modulation will be used on the link in the first phase, Costas loop is proposed for

demodulation because of ability to upgrade it to higher-level PSK demodulator.

Implementation was simulated in Matlab and Simulink, and implementation parameters are

described among results. We have also shown that it's possible to quickly implement

Matlab and Simulink implementation of demodulator in FPGA using Xilinx System

Generator, which was used for prototyping of the demodulator on Xilinx FPGA platform.

Third, a modular Ground station design was proposed on a logical level. Communication

circuit, which includes signal path to transceiver, and transceiver, was physically separated

from computers that can control communication circuit, and is software-defined (FPGA

demodulator and several software controlled components on transceiver with ARM

processor) and connected to server computers over the Ethernet network. Proposed design

is to be built from the ground up, with purpose of education about different components

function and design in Ground station (educational goals are part of the Trisat's goals), and

ability for easier demonstration of Ground station's and satellite's function for promotional

purposes of the Faculty, because communication with transceiver is IP-based. Therefore

multiple computers from the IP network on different locations can communicate with

transceiver station and therefore with satellite.

Further proposed work is implementation of not yet implemented components of the

Ground station, as well as development and launch of Trisat satellite, where link path

theory will be used in practice.


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APPENDIXES

A LINK BUDGET CALCULATIONS

A.1 GROUND STATION PART – DOWNLINK

Signal-121,7 -140

16,4 16,4-0,38 -0,38

20 20 0,470 70 1,5

-7,6 -7,6

-23,28 -41,58 1,916,4

Signal-132,3 -146

35,4 35,4Switch: -0,7 -0,7

-1,24 -1,2430 30

-2,97 -2,9730 30 225 25 0,5

-7,6 -7,6

-25,11 -38,81 211,7

UHF – downlinkNoise Noise Fig.

Input level:Antenna gain:Intermediate cable:LNA amplification:Additional gain necessary:Cables:Receiving system noise: -130 dBm (insignificant)Signal level at receiver input:Noise figure:

S-band downlinkNoise Noise Fig.

Input level:Antenna gain:

Intermediate cable:LNA amplification:Intermediate cable:Down converter:Additional gain necessary:Cables:Receiving system noise: -65 dBm (insignificant)Signal level at receiver input:Noise figure:


A.2 SYSTEM NOISE CALCULATIONS

Example system noise calculation. In our case, system noise is low enough not to play important role.

2,51E-0162,23279E-018

-146,51

S-band – downlink – noise receptionNoise level in antenna proximity:Noise level received by antenna [W]:in dBm:

1,00E-0168,22222E-018

-140,85

UHF – downlink – noise receptionNoise level in antenna proximity:Noise level, received by antenna [W]:in dBm:


System noise calculations, based on [27].

Transmission Line Coefficient: 1,0000

Antenna or "Sky" Temperature: NOTE: 4 K

Ground Station Feedline Temperature: 290 K

LNA Temperature: 290 K

LNA Gain: 40,0 dB 10000,0

Cable/Waveguide D Length: NOTE: 25,0 meteres

Cable/Waveguide D Type: Belden 9913 cable

Cable/Waveguide D Loss/meter: 0,092 dB/m

Cable/Waveguide D Loss: 11,0 dB

Communications Receiver Front End Temperature 290 K

System Noise Temperature: 294 KSystem Noise power: 2,03259E-010 W at BW of 50 MHzSystem Noise power: -65 dBm

Ground Station, Antenna or Sky Noise Temperature Calculation Tool:

Galactic Noise Component:

Receiver Frequency: 2400 MHz

Coldest Galactic Noise Temp.: 3 K

Warmest Galactic Noise Temp: 3 K

Terrestrial Noise Component:

Receiver Bandwidth: 50000,0 KHz

NOTE: Estimated or Measured Noise Level: -130 dBm

Noise Source Effective Temperature: 0 K

Minimum Sky Noise Temp: 3 K

Maximum Sky Noise Temp: 4 K

α =

Ta =

To =

TLNA =

GLNA =

TComRcvr =

Ts =


A.3 GROUND STATION PART – UPLINK

Signal18 -130

-5,8 -5,836 36

-0,29 -0,29

16,4 16,4

64,31 -100,09 0164,4

Signal18 -130

-7,6 -7,617 17

-2,97 -2,9731 0

-1,24 31Switch: -0,7 -0,7

35,4 35,43

85,92 -57,87 3

VHF – uplinkNoise Noise Fig.

Output level:Cables:Power amplifier:Intermediate cables:

Antenna gain:

Output level:Noise figure:

S-band uplinkNoise Noise Fig.

Output level:Cables:Upconverter:Intermediate cables:Power amplifier:Intermediate cables:

Antenna gain:

Output level:


A.4 SATELLITE PART

The following factors determine Path loss (other):

The main factor that influences loss of signals passing through Ionosphere, is signals

S-band(to)

21 306 6

-160 -160-0,6 -0,6-4,9 -4,9

-138,5 -129,5

(to)49 73

-160 -160-0,6 -0,6

6 6-4,9 -4,9

-110,5 -86,5

UHF&VHF(to)

27 330 0

-145,9 -145,9-1,3 -1,3-0,3 -0,3

-120,2 -114,2

(to)23 51

-136,4 -136,4-1,2 -1,2

0 0-0,2 -0,2

-114,6 -86,6

Satellite->Earth (from)TS output power:Antenna gain:Path loss (distance):Path loss (other):Pointing loss:Signal level on Earth:(Earth antenna gain: 35.4 dBi)

Earth->Sattelite (from)Earth station EIRP:Path loss (distance):Path loss (other):Antenna gain:Pointing loss:Input signal level:

Satellite->Earth (from)TS output power:Antenna gain:Path loss (distance):Path loss (other):Pointing loss:Signal level on Earth:(Earth antenna gain: 16.4 dBi)

Earth->Sattelite (from)Earth station EIRP:Path loss (distance):Path loss (other):Antenna gain:Pointing loss:Input signal level:(Expected in range between -120 dBm and -70 dBm)


frequency. Based on [27], the Ionospheric losses on VHF uplink are 0,4 dB, and on UHF

downlink around 0,8 dB. On S-band, expected Ionospheric losses are about 0,4 dB, based

on [49].

Atmospheric losses are mostly dependent on the elevation angle below frequencies of 2

GHz. In our case, with 45º elevation angle, determined loss is around 0,3 dB.

Also polarization loss is included here among other path losses. In our case, circular

polarization is used, so no loss is expected, but 0,2 dB of polarization loss is accounted

because of possible signal impairments that occurred during space travel of signals.

Pointing loss is determined based on our assumptions of pointing mismatch between

Ground station and Space station antennas. In UHF/VHF range, 5º mismatch is expected

on Ground station. Orientation of UHF/VHF antennas of Space station is not important,

because 0 dB attenuation and and 360º coverage is expected. Antenna pointing loss is

calculated 0,3 dB in UHF and 0,2 dB in VHF and is accounted in Path loss (other) in

UHF/VHF calculations. In S-band range, 2º pointing mismatch causes 4,9 dB of pointing

loss.


A.5 PATH LOSS CALCULATIONS

Pointing losses calculation:

Pointing losses calculation is shown on next page. Pointing loss is based antenna opening

angle and expected pointing mismatch with the following formula:poining loss in dB =-10*LOG10(3282,81*((SIN(RADIANS(PARAM))^2/(PARAM^2))))

where

PARAM = =2*(estimated_pointing_error*(79,76/INDEX(aperture_angle)))

= 159,5135,8

6.378,14 km 145,41.200,0 km300,0 km7.128,1 km

0,06313098,61180,0

7,134820,00

Period: 99,822-3,02051,0185750,00 km

7.128,14 km98,3345,0

1.009,94 km.

S-band: 2228,000 MHz 0,135VHF: 145,800 MHz 2,056UHF: 437,450 MHz 0,685

Path Loss = 22.0 + 20 log (S/λ) For S-band:For VHF:

Earth Radius: For UHF:Height of Apogee (ha):Height of Perigee (hp):Semi-Major Axis (a):Eccentricity (e): Inclination (I): degreesArgument of Perigee (ω): degreesR.A.A.N. (Ω): degreesMean Anomaly (M): degrees

minutesdω/dt: deg./daydΩ/dt: deg./dayMean Orbit Altitude:Mean Orbit Radius:Sun Synchronous Inclination: degreesElevation Angle (δ): degrees

Slant Range (S):

Frequency: Wavelength λ:metersmetersmeters



S-band:

2228 MHz 0,1346

4 18,5 3,2 °

2 ° 4,9

4 18,5 3,2 °

2 ° 4,9

7 6,0 85 °

2 ° 0,0

7 6,0 85 °

2 ° 0,0

UHF:

437,45 MHz 0,6853

3 2,0 180 °

5 ° 0,0

4 18,5 29,6 °

5 ° 0,3

VHF:

145,8 MHz 2,0562

3 2,0 180 °

Uplink: Downlink:

SpacecraftAntenna

[Types 1 thru 5]

Ground StationAntenna

[Type 1,2,3 or 4]

Parameters: Downlink Frequency: Wavelength: meters

Ground station downlinkAntenna:

User Defined Gain: dBiC Beamwidth:Result:

Esimated Pointing Error (θ4): Approx. Antenna Pointing Loss: dB

Ground station uplinkAntenna:



Space station downlink:Antenna:

Other (User Defined) [Isotropic Radiator] Gain: dBi Beamwidth:Result:

and vector from S/C to gnd. station (θ2): Pointing Loss: dB

Space station uplink:Antenna:

Other (User Defined) [Isotropic Radiator] Gain: dBi Beamwidth:Result:


Parameters:Downlink Frequency: Wavelength: meters

Space station:Antenna:

Canted Turnstyle Gain: dBiC Beamwidth:Result:


Ground station:Antenna:



Parameters:Uplink Frequency: Wavelength: meters

Space station:Antenna:

Canted Turnstyle Gain: dBiC Beamwidth:Result:

θ2

θ1θ3

θ4

+Z

+X-Z

+Z

+X -Z

Spacecraft Symmetry Axis


B XILINX SYSTEM GENERATOR PROTOTYPE OF DEMODULATOR

B.1 SYSTEM GENERATOR MODEL SPECIFICATIONS

• Input sampling period: 0.000001 s

Input sampling frequency: 1000 kHz = 1 MHz

• Signal sampling precision: 16-bit

• FPGA clock period : 10 ns

FPGA clock frequency: 100 MHz

• DDS Compiler system clock: 10 MHz

• DDS required frequency at startup: 100 kHz

• Simulink system period: 0.000001 s

Simulink simulation frequency: 1000 kHz = 1 MHz

B.2 SYSTEM GENERATOR MODEL

The Xilinx System Generator model is depicted in consists basically of the following

components:

• Input signal, imported from Matlab workspace. It is the input data, produced by

pure Matlab model of modulator, channel, downconversion and demodulator,

before at the point after downconversion and before reaching demodulator. This

data is saved into file for usage with Xilinx System Generator design of

demodulator.


• Input data, which is in synchronism with input signal for the propose of

monitoring the function of demodulator.

• Model of demodulator, created with Xilinx System Generator blocks, which is

described in the following section.

• Scope for observation of demodulation results

The model on top level is shown on Figure B.1.

B.3 SYSTEM GENERATOR MODEL OF DEMODULATOR

This model is an implementation of Matlab demodulator, using only Xilinx System

Generator blocks, and is shown on Figure B.1. By using only blocks, provided by System

Generator, we can target and compile the design for particular hardware platform and load

it on FPGA, therefore achieving a working hardware implementation of the design. In the

hardware implementation, some additional parameters, like bit width of interfaces between

blocks and clocking of elements must be taken into consideration. Also the Matlab

Figure B.1: System Generator model on top level.


simulation period must be set accordingly to frequencies at which the system is going to be

clocked in order to achieve useful simulation results.

The borders of the model which can be compiled to hardware level are set by In and Out

gateway blocks. In our system, there is one In gateway block, which receives the signal

downconverter, therefore the signal that is input into our Xilinx System Generator model's

demodulator block.

That signal is then multiplied with the sine and cosine wave from NCO, which is described

in B.4 System Generator model of NCO and passed through "arm filters". The filters are

realized using Xilinx FIR Compiler blocks and use coefficients, which are the same as the

coefficients in Matlab design. Because of the hardware limitations, only filters with up to

48 coefficients are used.

B.4 SYSTEM GENERATOR MODEL OF NCO

The NCO, shown on Figure B.2, which is fed by the rest of the loop and provides sine and

Figure B.2: Costas loop model using only Xilinx System Generator blocks


cosine reference signal to the loop, is implemented using Xilinx DDS Compiler block for

Direct Digital Synthesis. This block can be configured with different options and basically

consists of Phase generator and Sinus and Cosinus lookup tables. We use all of the

components. DDS Compiler block allows different options of programming its output

frequency thorugh programming phase increment (which is used at every clock cycle to

increment the pointer and point to the next value in the lookup tables) and phase shift.

Throught chaning of phase increment, DDS block can be preprogrammed to output certain

frequency, the frequency could be fed in with particular clock cycles (programmable), or it

could be programmed in a streaming way.

Because of the nature of our application, which dynamically adjusts output frequency of

the NCO, the phase increment is programmed in a streaming way. On the input, the DDS

Compiler block always expects a value between 0 and 1, where 1 is the phase increment of

the whole cycle. The phase increment input (pinc_in) bit width determines how fine could

be the frequency adaptations.

To achieve the initial frequency of the NCO (which is the frequency at which we expect

the carrier signal), we use an initialized register with value of phase increment, calculated

with the following equation from [47]:

increment=( f des∗2B)

f clk

(App.1)

when streaming input, this increment must be provided to DDS Compiler core in form:

input= increment2B

(App.2)

where B is the DDS Compiler input bit width, fdes is desired output frequency and fclk is the

clock frequency of DDS Compiler component.

The initial value is then adjusted, according to the NCO block input, in the following way:


1. the input is between -1 and 1, and is first multiplied by constant, which is the

smallest number that can be written by the amount of bits, used by pinc_in. In that

way, the input frequency change value is diminished to the level at which it may

affect the pinc_in value. The output of multiplication must have double the bit

width of the constant, because the output numbers are always smaller than the

smallest number which could be written by the bit width of the constant. The output

is signed.

2. the signed change of the phase increment is then added to the previous value of

phase increment.

3. the new phase increment is then fed into DDS Compiler block. After addition, the

value is always positive and is fed to a register for the next cycle.

4. In every cycle, the value of pinc_in is fed from register to the DDS Compiler block.

Figure App.3: System Generator NCO model.




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