Phased Array Antenna Element for Automotive Radar

Application

by

Mohammad Hossein Nemati

Submitted to the Graduate School of Engineering and Natural Sciences

in partial fulfillment of

the requirements for the degree of

Master of Science

Sabancı University

Summer, 2014

ii

Phased Array Antenna Element for Automotive Radar Application

APPROVED BY

Prof. Dr. İbrahim Tekin

(Thesis Supervisor)

……………………………………………

Assoc. Prof. Dr. Ayhan Bozkurt

……………………………………………

Assoc. Prof. Dr. Melih Papila

……………………………………………

DATE OF APPROVAL: ……………………………………………

iii

© Mohammad Hossein Nemati 2014

All Rights Reserved

iv

Acknowledgements

I would like to offer special thanks to my supervisor Prof. Ibrahim Tekin for his help and

consideration through the completion of my master study. I also like to thanks my

colleagues at Sabanci University.

v

Phased Array Antenna Element for Automotive Radar Application

Keywords: Phased Array Radar, Patch Antenna, Surface Wave, W-band Dielectric

Measurement, SiGe BiCMOS,

Abstract

In this thesis work, a design of reliable antenna front-end for W band automotive radar is

studied and the problems and considerations associated with phased array antenna design at

W-band are addressed. Proposed phased array antenna consists of on chip patch antenna

which has the advantages of being integrated by the active circuitry. A sample of patch

antenna and patch array are designed and fabricated to be tested for their functionality.

Printing antenna on Silicon substrate is a compact and cost-effective approach. However,

antenna on Silicon will have poor gain and will also suffer from surface wave (SW)

excitation. The reason for this is Silicon high dielectric constant and loss. Surface wave

can be easily excited on high dielectric constant substrate which results in gain drop and

distortion in radiation pattern. To avoid substrate loss, available back etching in foundry

process is used to remove the silicon under the radiating antenna and improve the gain. To

kill the surface wave, a type of engineered material- called Electromagnetic Band-Gap

(EBG)- is designed to filter the SW around the antenna's frequency of operation. To test the

fabricated antenna, a measurement setup is implemented to do refection coefficient and

radiation pattern measurement. Measured S parameters show that there is frequency shift in

response of measured antenna with respect to the simulated one. This shift can be attributed

to uncertainty about the dielectric constant of Silicon at W-band. To find the exact value of

Silicon dielectric constant, a measurement setup based on free space method is devised to

determine the exact value of the silicon dielectric constant at W-Band frequency range.

vi

Otomotiv Radar Uygulamaları için Faz Dizinli Anten Elemanı

Anahtar Kelimeler: Faz Dizinli Radar, Yama Anten, Yüzey Dalgası, W-bant Dielektrik

Ölçümü, SiGe BiCMOS

Özet

Bu tezde, W-bant otomotiv radar için güvenilir bir ön-uç anten tasarımı anlatılmış ve W-

bant’ta faz dizinli anten tasarımlarındaki sorunlara ve önemli noktalara değinilmiştir.

Önerilen faz dizinli anten, aktif devrelerle entegre edilebilme avantajına sahip olan kırmık

üstünde yama antenden oluşmaktadır. İşlevselliği test etmek üzere, yama anten ve anten

dizini örnekeleri tasarlanmış ve üretilmiştir. Antenin silikon substrat üzerine basılması

kompakt ve masrafızdır. Ancak, silikon üzerindeki antenin kazancı düşük olmakla birlikte,

aynı zamanda yüzey dalgalarından (SW) da etkilenmektedir. Bunun nedeni, silikonun

yüksek dielektrik sabiti ve kaybıdır. Yüzey dalgaları, kazancın düşmesine ve radyasyon

paterninin bozulmasına sebep olan yüksek dielektrik sabitli substratlarda kolayca oluşabilir.

Bu kayıplardan kurtulmak için, ışıma yapan antenin altındaki silikon, üretim sürceinde ters-

aşındırma ile çıkarılmış ve kazanç artırılmıştır. Yüzey dalgalarından kurtulmak için ise,

antenin çalışma frekansı etrafındaki SW’yi filtreleyen özel bir Elektromanyetik Bant-

Açıklığı (EBG) tasarlanmıştır. Üretilen anteni test etmek için, yansıma sabiti ve radyasyon

paterni ölçen bir düzenek kurulmuştur. Ölçülen S-parametreleri incelendiğinde,

simulasyonlardakine kıyasla, frekans kayması gözlenmiştir. Bu sapma, W-bant’taki

silikonun dielektrik sabitinin belirsizliğinden kaynaklanmaktadır. W-bant’ta silikonun

dielektrik sabitinin kesin değerini bulmak için, free-space yöntemine dayanan bir ölçüm

düzeneği kurulmuştur.

vii

Contents

ABSTRACT ........................................................................................................................... V

ÖZET ................................................................................................................................... VI

LIST OF FIGURES ........................................................................................................... VIII

1. INTRODUCTION ........................................................................................................... 1

2. RADAR SYSTEM REQUIREMENTS AND CHOICE OF ANTENNA ...................... 5

2.1 ANTENNA CHOICE FOR 77GHZ SHORT RANGE RADAR SYSTEM .................................. 9

2.2 PATCH ANTENNA'S WORKING PRINCIPLE ................................................................. 10

2.2.1 Feeding Method of Patch Antenna ..................................................................... 14

3. PATCH ANTENNA DESIGN FOR 77GHZ RADAR SYSTEM ................................ 19

3.1 SOLUTION FOR SURFACE WAVE PROBLEM .............................................................. 26

3.1.1 Surface wave: ..................................................................................................... 27

3.1.2 Application of engineered material to filter surface wave ................................. 29

3.2 ELECTROMAGNETIC BAND-GAP META-MATERIAL: ................................................. 30

3.2.1 Periodic holes: .................................................................................................... 31

3.2.2 Bumpy Surfaces .................................................................................................. 32

3.2.3 Uni-planar Compact photonic Ban-dgap (UC-PBG): ........................................ 33

3.3 CHARACTERIZATION OF THE 2-D UNI-EBG STRUCTURE: ........................................ 35

3.4 EBG-PATCH ANTENNA DESIGN: ............................................................................. 39

3.5 ANTENNA INTEGRATION WITH ACTIVE CIRCUITRY .................................................. 42

viii

4. ANTENNA FABRICATION & MEASUREMENT RESULT .................................... 44

4.1 SIMULATED AND MEASURED RESULT: ..................................................................... 46

4.2 DIELECTRIC CONSTANT MEASUREMENT: ................................................................. 48

4.2.1 Theory of dielectric constant derivation ............................................................. 49

CONCLUSION AND FUTURE WORK ............................................................................. 56

REFERENCE ........................................................................................................................ 59

List of Figures

Fig. 2.1 RF Front End of Automotive Radar System ............................................................ 8

Fig. 2.2 Geometry of Patch Antenna (Top & Side View) .................................................. 11

Fig. 2.3 Normalized Radiation Pattern for Microstrip (Patch) Antenna (a) E plane, (b)

Hplane ........................................................................................................................... 12

Fig. 2.4 Side View of patch antenna with E-fields Shown Underneath .............................. 13

Fig. 2.5 Patch Antenna with an Inset Feed .......................................................................... 15

Fig. 2.6 Patch antenna with a quarter-wavelength matching section ................................... 16

Fig. 2.7 Coaxial Cable Feed of Patch Antenna .................................................................... 17

ix

Fig. 2.8 Coupled (indirect) Inset Feed ................................................................................. 17

Fig. 2.9 Aperture Coupled Feed ........................................................................................... 18

Fig. 3.1 Different pattern of Substrate Etching .................................................................... 20

Fig. 3.2 Patch Antennas over Etched Silicon Substrate ....................................................... 21

Fig. 3.3 Radiation pattern on E and H plane ........................................................................ 21

Fig. 3.4 Return Loss and Mutual Coupling of Array of two Patches .................................. 23

Fig. 3.5 Beam Rotation at H-Plane ..................................................................................... 24

Fig. 3.6 Radiation pattern change at (a) Eplane (b) Hplane .............................................. 25

Fig. 3.7 Radiation Pattern (a)Patch with Small Substrate & (b) Patch with Large Substrate

...................................................................................................................................... 26

Fig. 3.8 (a) dielectric coated conductor (b) Electric field(solid line) and magnetic field

distribution of TM0 mode (c) Patch antenna first propagating mode ........................... 28

Fig. 3.9 Microstrip Transmission line with/without EBG ................................................... 32

Fig. 3.10 Mushroom EBG structure ..................................................................................... 33

Fig. 3.11 Tripole Metallodielectric Photonic Band-gap (MPBG) and Uni-EBG structure . 34

Fig. 3.12 Uni-planar Compact EBG Structure Which realizes a 2D Periodic Network of LC

Circuits without Introducing Vias ................................................................................. 34

Fig. 3.13 Uni-planar EBG Surface and HFSS Model for unit cell of Uni-Planar EBG

Surface .......................................................................................................................... 35

Fig. 3.14 Surface Wave Dispersion Diagram for EBG Structure ........................................ 36

Fig. 3.15 Surface Wave Dispersion Diagram for EBG Structure with Doubled Size ........ 37

Fig. 3.16 Two Different Modeling Scheme of EBG to Derive S Parameter ....................... 38

x

Fig. 3.17 S11 & S21 for (a) Ideal Microstrip Transmission Line (b) Line Surrounded with

EBG Substrate ............................................................................................................... 39

Fig. 3.18 Dispersion Diagram for EBG with Band-gap around 77GHz .............................. 40

Fig. 3.19 Designed Antenna with its S11 ............................................................................ 41

Fig. 3.20 Radiation pattern for two type of antenna ............................................................ 42

Fig. 4.1 Schematic of Millimeter Wave Antenna Setup ...................................................... 45

Fig. 4.2 Implemented Millimeter Wave Antenna Setup ...................................................... 46

Fig. 4.3 Simulated and Measured S11 for dipole antenna from previous work .................. 47

Fig. 4.4 Schematic of the Measurement Setup ................................................................... 50

Fig. 4.5 Implemented Dielectric Setup ................................................................................ 53

Fig. 4.6 Measured Phase Difference versus Frequency ....................................................... 54

Fig. 4.7 Calculated dielectric constant ................................................................................. 55

1

1. Introduction

There are increasing application areas for automotive radar devices in cars all around

the world. These devices are employed in advanced cruise control systems, which can

actuate a motor vehicle’s accelerator and/or brakes to control its distance separation

behind another vehicle. It is anticipated that the use of these systems will become

commonplace in the future, especially the civilian use of millimeter wave radars in

navigation, road traffic control, safety for highway driving and security [1]. There are

two bands allocated for automotive radar applications: 24 and 77-GHz band. The 24-

GHz band consist of, one around 24.125GHz with BW of 200MHz, another 24GHz

with BW of 5GHz for short/mid-range radars. In 77 GHz band, there is a 5 GHz

bandwidth (76-77-GHz for long-range adaptive cruise control and 77-81-GHz short-

range radar sensors) allocated for application of short range automotive radar for the

purposes of stop and go, blind side detection, crash avoidance. Radar range resolution is

inversely proportional to available bandwidth (2 GHz bandwidth corresponds to a 15

cm range resolution). As the bandwidths are much larger in millimeter wave frequencies

compared to lower frequencies, for a 4-5 meter size car, 15 cm range resolution seems

to be enough for detection purposes [2]. Resolution can be easily handled by the

2

already available wide bandwidth of 2 GHz at 77 GHz. RF-front end of radar system

comprises of array of antenna followed by phase shift mechanism to make beam

steering possible. The choice of antenna will be dependent on the radar requirements

such as max distance and cost and ease of fabrication.

At 77 GHz, the size of the antenna becomes comparable to the chip size (less than 1

mm), and this creates the opportunity for a single-chip transceiver to be integrated with

the antenna or antenna arrays. There is extensive research on designing on-chip

antennas for 77 GHz band including unbalanced type of antennas such as slot and

microstrip patch type antennas as well as balun circuits feeding balanced antennas like

dipoles [3]. Planar type antenna like Microstrip patch antennas are good candidate for

this purpose due to the advantages of low profile, light weight, and low cost [4]. So,

SiGe foundry process can be used to realize overall radar system circuitry and make

fabrication easy and more cost effective. However, there would be challenges for

printing antenna on high dielectric constant substrate like Silicon such as gain drop and

performance degradation due to unwanted surface wave and Silicon loss. In order to

tackle the problems caused by surface wave in patch antenna, two approaches have been

pursued to have a patch antenna with optimum performance on high-dielectric constant

substrate. First method uses micromachining technology, while the second makes use of

the concept of electromagnetic band-gap (EBG) structures [5]. In the first approach, part

of the substrate right under the radiating element is removed to establish a low effective

dielectric-constant environment for the antenna. Doing so, power loss due to surface-

wave excitation is reduced and efficiency of energy coupling to space waves improves.

The second approach take advantages of EBG structures: the high-permittivity substrate

is engineered by putting periodic structure to change the propagation characteristic of a

surface wave around antenna operative frequency. Various types of periodic loading of

substrate have been studied [6]. One method is to drill a periodic pattern of holes in the

substrate or ground plane. Another method is to embed a periodic pattern of metallic

pads inside the substrate; pads are shorted to the ground plane with vias. In the last one,

a type of planar or 2-D loading (no vias are required) were proposed which is

compatible with RFIC integrated circuit fabrication technology (uni-planar

electromagnetic bandgap (Uni-EBG)). This thesis work will mostly be about developing

3

reliable antenna system for 77GHz radar system and will address the challenges ahead

of designing and fabricating antenna at W-band.

There is uncertainty about the dielectric constant of substrates such as silicon at W-band

frequencies. Silicon dielectric uncertainty effect becomes very important for the design

and simulations of RF circuits and antennas on silicon substrates. Designing an antenna

on a substrate with inaccurate value of dielectric constant will change antenna

performance and shift the resonance frequency in practice which causes large

discrepancies between simulated and measured results. Simple and applicable methods

for measuring the permittivity of the substrate are always in great interest of the

microwave circuit and antenna designers. There are various methods for measuring

dielectric constant including dielectric waveguide, cavity resonator, open resonator and

free space [7]. Free space method is more suitable for wideband measurement. In free

space method, the dielectric constant is measured based on either the transmission

method or metal backed method [8-9]. This thesis also tries to address this issue. For

this, two approaches are applied to bring out the real part of silicon dielectric constant

based on transmission and reflection type measurement method. Only the phase

information of the transmitted signal is used to determine the real part of the

permittivity.

The thesis is organized as follow:

Chapter 2 studies the automotive radar system requirement regarding antenna and phase

shifting mechanism and discusses different antenna type that can meet the radar system

demands.

Chapter 3 presents the design of sample patch antenna on silicon substrate together with

discussion of methods that can overcome the challenges of designing antenna on Silicon

substrate.

Chapter 3 is about implementation of measurement setup at W band for antenna and

dielectric constant measurement. Measured and simulated results are also presented in

this chapter.

4

Chapter 4 presents the design of multiport single element antenna which is a parallel

research study with this work. Designed antenna could have potential application in

MIMO system. Finally, chapter five present the conclusion and future work in line with

this work.

5

2. RADAR SYSTEM REQUIREMENTS AND

CHOICE OF ANTENNA

Radar Technologies have been in use for defense purposes since World War II. With the

advance of the solid state technology (CMOS circuits up to 100 GHZ, SiGe Circuits

reaching to almost 1 THz), we see more civilian use of millimeter wave radar especially in

safe highway driving, navigation and traffic control. In line with these developments, ETSI

has developed standards for ‘’short range’’ radar for automotive applications in 24 – 77

GHz bands. In 77 GHZ band, there is a 2 GHz bandwidth allocated for an application of

short range automotive radar for the purposes of stop and go, blind side detection, crash

avoidance, braking if crash cannot be avoided and to keep safe driving distance with the

traffic ahead [1].

Radar range resolution is inversely proportional to available bandwidth (2 GHz bandwidth

corresponds to a 15 cm range resolution). As the bandwidths are much larger in millimeter

6

wave frequencies compared to lower frequencies, for a 4-5 meter size car, 15 cm range

resolution seems to be enough for detection purposes [2]. Resolution can be easily handled

by the already available wide bandwidth of 2 GHz at 77 GHz. One of the critical issues at

77 GHz band and beyond is the low received signal levels. This is due to both the limited

power output of the solid state power amplifiers (10-15 dBm) and also the high free space

loss due to small wavelengths (at 77 GHz, at 1-meter, free space loss is -70 dB). To

increase the received signal level at the receiver, one can use conventional antenna

technologies such as high gain phased array design, better design of single element

antennas and/or higher gain low noise amplifier. Note that at these frequencies, wavelength

is on the order of a few mms, and all the system including antennas can be made on a

single-chip. However, the silicon substrate for such an integrated system becomes very

lossy and for the case of antennas, care should be given to design of these antennas on-chip.

Radar operating range could be doubled or tripled depending on the gain of the phased

array system.

Antenna gain is primary important factor which determines the range of radar system and

SNR requirement. The higher the gain the higher would be the detectable range and SNR.

Hand calculation can be done, based on Friis formula (equation 2.1), to determine the range

and SNR for specific value of gain. For a radar system with BW=1GHz at 77GHz, the SNR

can be calculated for 5m range as follow. Using antenna in array can increase the gain and

result in improved SNR. Table.1 shows the link SNR budget for this scenario.

7

GHzBW

KT

ek

N

PSNR

BWTkN

RGPP

GRR

GPP

r

tr

tr

1

300

231*38.1

**

1

4

1

44

1

4

1

2

43

2

222

(2.1)

Table 1. SNR budget calculation

Single antenna SNR Array antenna SNR

dBSNR

GHzBW

GHzf

m

mR

dBiG

dBmPt

1

1

77

1

5

2

20

2

dBSNR

GHzBW

GHzf

m

mR

dBiG

dBmPt

11

1

77

1

5

8

20

2

Figure 1 shows the overall topology of proposed phased array system. The radar system

consists of array 4x1 antenna followed by active circuitry; the number of antenna can be

increased in array for more gain. Active circuitry consists of W band LNAs and phase

shifter with power combiner.

8

Fig. ‎2.1 RF Front End of Automotive Radar System

The antenna elements that will be used in the design could be a dipole, a bent-dipole and

Microstrip patch antenna or even planar slot type antenna. The choice of antenna will be

dependent on radar range and the assembly consideration. The output of the antenna will be

amplified by a W band two stage LNA circuit. For the antenna pattern steering, a 4x1 array

will be designed and each antenna line will have a very low loss MEMS based phase

shifter. Using these MEMS phase shifters, the antenna main beam will be steerable. Also,

one can appreciate the very low loss phase shifters, since at these frequencies; signal levels

are already too low. At these frequencies, MEMS based phase shifter is preferred due to

their low loss. Using IHP MEMS technology, it is already shown that low loss around 0.5

dB can be obtained. After the antenna, LNA and the phase shifter, a corporate type power

combiner will be used to combine the signals coming from the four antennas. All the

antennas, phase shifters, combiner and the LNA will be on the same single chip.

9

2.1 Antenna choice for 77GHz short range radar system

There is extensive research on designing on-chip antennas for 77GHz band including

unbalanced type of antennas such as slot and microstrip patch type antennas as well as

balun circuits feeding balanced antennas like dipoles [3]. The explorations and publications

on various types of on-chip antennas for upper 60 GHz band show very low or negative

gain without incorporating them with gain enhancement method like using reflectors. So,

Conventional design of these antennas on a Silicon substrate at 77 GHz band will not

generate antennas with much gain due to high silicon substrate loss and also deteriorating

effect due to Surface wave. Small size of the manufactured device and the substrate effect

are basic challenges and derives the interest in the methods of feeding, measuring, design

and fabrication of these types of antennas. However, in our project using IHP integrated

MEMS technology, substrate will be etched and since this effect is considered for the

proper design of the antenna, much higher antenna gains can be obtained. Available etching

technology will make it possible to remove the lossy Silicon substrate right under the

radiating antenna and create low profile substrate for antenna. For surface wave undesired

effects a filtering structure can be used to omit the SW around the frequency of interest.

The choice of antenna will be dependent on the radar range requirement and fabrication and

assembly consideration. Based on the calculation done in previous section using Friss

Formula (equation 2.1), an antenna with moderate gain can fulfill the gain requirement of

short range radar. Moreover, planar type antennas are attractive due to their low profile,

low cost and ease of integration. Considering these two factors, the antenna elements that

will be used for 77GHz short range system could be a dipole and microstrip patch antenna

or even planar slot type antenna. To increase the antenna gains further in dipole case, a

metal ground can be placed on one side of the dipole and bent dipole antennas to have gain

around 4-5dB. For single patch antenna a gain of 8dB can be obtained. Further gain

increase can be obtained by using antenna in array configuration. In this work, design of on

chip patch antenna is studied. Next section will go through the basic principle of patch

antenna.

10

2.2 Patch antenna's working principle

Microstrip patch antennas are widely used in wireless communications due to the

advantages of low profile, light weight, and low cost. Recent applications have pushed the

frequency into the mm-wave region for application such as automotive radars at the 77

GHz band [4]. Design of patch antenna is straight forward and there are closed form

formula based on the frequency of operation and the type of substrate [10].

Figure 2.2 shows a microstrip patch antenna which is connected to main feed line by mean

of microstrip transmission line. The antenna is printed on a substrate of thickness h with

permittivity εr. The other side of the substrate is ground plane with dimension larger than

that of patch. Both the patch and the ground plane are made of high conductivity metal and

their thickness is not important. As shown in figure, the patch length is L, width W.

Normally, the height of substrate h is should be a small fraction of wavelength of

operation(λ), but not much smaller than 0.05λ. However, in designing patch antenna on

high dielectric constant substrate like silicon care should be taken to keep the h as low as

possible to couple less energy to unwanted mode inside the substrate. This unwanted inside

the substrate-called surface wave- can degrade the antenna gain, radiation pattern and cross

polarization.

11

Fig. ‎2.2 Geometry of Patch Antenna (Top & Side View)

The length L of patch antenna determines the frequency of operation and the center

frequency approximately can be expressed as:

(‎2.1)

Equation (2.1) shows that the microstrip antenna should have a length equal to one half of a

guided wavelength. Guided wavelength is the wavelength inside the substrate which is

smaller than that of air. The width W of the microstrip antenna determine the antenna input

impedance and also the antenna bandwidth. Typically, input impedance for square patch is

around 300 Ohms. Increasing the width of antenna, the impedance can be reduced and

bandwidth can be increased. However, increasing the patch width will make the antenna

bulky which occupy a lot of space. The width further has effect on antenna's radiation

pattern. The normalized radiation pattern approximately can be expressed as:

( 2.2)

(‎2.3)

12

In the above, k is the free-space wave number, given by

. The magnitude of the fields,

given by:

( ‎2.4)

Typical radiation pattern of the microstrip antenna is shown in Figure 2.3.

Fig. ‎2.3 Normalized Radiation Pattern for Microstrip (Patch) Antenna (a) E plane, (b)

Hplane

In Figure 2.3, θ is angle from antenna broadside and the pattern is shown for E & H plane.

Patch antennas have directivity around 5-7 dB. The antenna is linearly polarized and it is

possible to have circular polarization by changing the feeding mechanism of the antenna.

Primary drawback of patch antenna is its small bandwidth and these types of antenna are

narrowband; the bandwidth of rectangular microstrip antennas is around 3%. However, the

patch antenna can be modified in a way to have larger bandwidth. The reason why the

patch antenna is radiating is because of fringing field around the antenna. Figure 2.4 shows

the side view of a patch antenna. Considering the current distribution on patch, it would be

zero at the ends (open circuit ends) and the maximum at the center of half-wave patch. So,

the current at patch input is almost zero and this low current value at the feed explains high

value of impedance at patch input. Another interpretation is based on voltage distribution

on patch antenna. Considering patch as open circuited transmission line, the voltage

13

reflection coefficient will be -1. This means that the current and voltage distribution are out

of phase with respect to each other. Therefore, the voltage is maximum (say +V volts) at

patch end and is minimum at the start of patch (-V Volts). By this expression, the fields

underneath the patch will looks like that of Figure 2.4, which shows the fringing of the

fields around the edges.

Fig. ‎2.4 Side View of patch antenna with E-fields Shown Underneath

Fringing fields at the patch make patch to radiate the EM wave. As shown in Figure 2.4,

the fringing fields near the surface of the patch antenna are both in the +y direction. So, the

fringing E-fields on the edge of the microstrip antenna are at the same phase and add up in

phase and produce the radiation of the microstrip antenna. To understand the radiation

mechanism of patch it's important to know the fringing fields. The current distribution on

patch cannot describe the radiation mechanism since the current distribution on patch is

opposite of current distribution on ground which cancel each other. However, this can

explain why microstrip transmission line does not radiate. The microstrip antenna's

radiation is due to the fringing fields, which are due to the advantageous voltage

distribution; hence the radiation arises due to the voltage and not the current. So, the patch

antenna can be called "voltage radiator", which is different from wire antenna that radiate

due to advantageous current distribution. In wire antennas, the currents add up in phase and

are therefore "current radiators". An example of current radiator is dipole antenna.

14

It also worthwhile to mention that the smaller is, the fringing fields will extend farther

away from the patch and make the patch better radiator. In other word, higher dielectric

constant will more confine the electric field inside the substrate and decrease the radiation.

Therefore, using a smaller permittivity for the substrate yields better radiation. In contrast,

for microstrip transmission line, it's desired to have as less as radiation. So, a high value of

is desirable for feed line to have less fringing filed and less radiation This is one of the

trade-offs in patch antenna design. It is possible to use substrate with different permittivity

for feed line section and patch antenna to improve patch radiation property and reduce that

of transmission line.

2.2.1 Feeding method of patch antenna

Patch antenna can be fed in different ways: Based on the application an appropriate feeding

method can be applied.

Inset Feed: as described earlier, patch antenna has high input impedance due to low current

at the end . It is possible to modify the feed and match the antenna impedance to

that of 50 Ohms line. The current is low at the open ends of a half-wave patch and increases

in magnitude toward the center, the input impedance could be reduced if the

patch was fed closer to the center. One method of doing this is by using an inset feed (a

distance R from the end) as shown in Figure 2.5.

15

Fig. ‎2.5 Patch Antenna with an Inset Feed

Current has a sinusoidal distribution and moving in a distance R from the end will increase

the current by

- this is just noting that the wavelength is , and so the phase

difference is

. The voltage also decreases in magnitude by the same amount that

the current increases. Hence, using , the input impedance scales as:

( ‎2.5)

In the above equation, Zin(0) is the input impedance if the patch was fed at the end. Hence,

by feeding the patch antenna as shown, the input impedance can be decreased. For

example, if , then

, so that . So, a (1/8)-

wavelength inset would decrease the input impedance by 50%. This method can be used to

have desired value for input impedance which is 50 Ohms in normal case.

Fed with a Quarter-Wavelength Transmission Line: the microstrip antenna can also be

matched to a transmission line of characteristic impedance Z0 by using a quarter-

wavelength transmission line of characteristic impedance Z1 as shown in Figure 2.6.

16

Fig. ‎2.6 Patch antenna with a quarter-wavelength matching section

The goal is to match the input impedance (Zin) to the transmission line (Z0). If the

impedance of the antenna is ZA, then the input impedance viewed from the beginning of the

quarter-wavelength line becomes

( ‎2.6)

This input impedance Zin, can be equated to Z0 (Zin=Z0) for specific value of Z1 and the

antenna is impedance matched. The impedance Z1 can be tuned by changing the width of

the quarter-wavelength strip. To lower the characteristic impedance (Z0) we need to make

the width of strip larger.

Coaxial Cable or Probe Feed: Microstrip antennas can also be fed by coaxial cable using

via that connects the patch to center conductor of patch as shown in Figure 2.7. The outer

conductor of the coaxial cable is connected to the ground plane in other word is grounded.

17

Fig. ‎2.7 Coaxial Cable Feed of Patch Antenna

The position of the feed can be changed to have 50 Ohm input impedance like the case of

inset feed. Care should be taken about the inductance introduce by the coaxial feed. For this

a height h of substrate should be considered to be low to neglect the inductance effect. In

addition, the probe will also radiate, which can lead to radiation in undesirable directions.

Coupled (Indirect) Feeds: This is non contacting feeding method which is the modified

version of inset feed. The inset feed can also be stopped just before the patch antenna, as

shown in Figure 2.8. Moreovere, the probe feed in Figure 2.5 can be trimmed such that it

does not extend all the way up to the antenna.

Fig. ‎2.8 Coupled (indirect) Inset Feed

18

The advantage of the coupled feed is that it adds an extra degree of freedom to the design.

The gap introduces a capacitance into the feed that can cancel out the inductance added by

the probe feed.

Aperture Feeds: Another method of feeding microstrip antennas is the aperture feed. In

this technique, the feed circuitry (transmission line) is shielded from the antenna by a

conducting plane with a hole (aperture) to transmit energy to the antenna, as shown in

Figure 2.9.

Fig. ‎2.9 Aperture Coupled Feed

This method is good to make patch to radiate efficiently and make the transmission line

radiation less. For this purpose, the upper substrate can be made with a lower permittivity

to produce more extended fringing fields, yielding better radiation. The lower substrate can

be independently made with a high value of permittivity to have more confined field. The

disadvantage of this method is increased difficulty in fabrication.

19

3. Patch Antenna Design for 77GHz Radar

System

This chapter will discuss design of patch antenna at 77GHz and the challenges ahead for

designing antenna on high dielectric constant substrate like Silicon. As discussed in

previous chapter, design of patch antenna is straightforward and there are analytical

formulas for this purpose. However, this formula can be used for patch antenna on

homogeneous substrate not etched substrate; substrate etching will be used to have low

profile substrate and improve antenna gain. To design antenna on etched substrate, effective

dielectric constant should be calculated before using analytical formulas and it is not easy

to find closed form formula for this purpose. However, it's obvious that the effective

dielectric constant would be less than that of Silicon. To begin with, dielectric constant can

be consider 1/8 of Silicon, then full EM simulator like HFSS can be used to numerically

model the problem. Tuning the parameter, the desired resonance frequency can be obtained

at 77GHz with specific value of gain.

20

The best way to do the etching is to remove as much as Silicon underneath the radiating

element to have regarding the available etching technology. IHP SiGe technology allows

rectangular shape etching with max 700x700mm and minimum of 100x100mm. Figure 3.1

shows different etching pattern tried for antenna with maximum gain. Simulation shows

that the last etching pattern is effective one regarding the gain and overall size.

Fig. ‎3.1 Different pattern of Substrate Etching

Figure 3.2 shows a simple patch antenna on etched silicon substrate based on the MMIC

technology available at IHP with etching pattern shown in Figure 3.1d. Feeding

mechanism is chosen to be inset fed in order to save more area. Overall size of the antenna

is 2mm2. Designed antenna's resonance frequency is at 77GHz and the 10dB bandwidth is

about 5GHz from 75-80GHz.

21

Fig. ‎3.2 Patch Antennas over Etched Silicon Substrate

Figure 3.3 shows the radiation pattern on E and H plane.

Fig. ‎3.3 Radiation pattern on E and H plane

5GHz Bandwidth

22

Printing patch antenna on low loss and low dielectric constant substrate like FR4, the

antenna will have gain of around 8dBi at broadside and undistorted radiation pattern both at

E & H plane. However, using Silicon as antenna substrate will result in 2-3dB gain drop

and distorted pattern especially in Eplane. This effect can be attributed to unwanted modes

of propagation inside the high dielectric substrates called surface wave. The higher the

dielectric constant, the higher would be the possibility of surface wave excitation.

Moreover, larger height of substrate will increase the possibility of SW excitation. SW

results in gain drop, pattern distortion and high cross-polarization. Moreover, the amount of

energy coupled to SW increase with increase in height of substrate. So, the substrate height

should be kept as low as possible. Surface wave issue will be addressed elaborately in

upcoming section.

For the time being, we will focus on designing antenna system that meets requirements of

phased array system. Primary requirements are high gain and beam steering capability.

Designed single patch has realized gain of 5dB. To increase the radar antenna front end

gain and also make the beam steering possible, array configuration can be used. To verify

the concept, array of two patch antennas were designed. The beam rotation would be in

patch H plane since it makes feeding network simple and also benefits from undistorted

pattern at Hplane. To increase the gain more, the number of element should be increased in

array. Figure 3.4 shows return loss and mutual coupling of array with two patch elements.

Mutual coupling is better than -15dB inside the band and can be further improved by

increasing the spacing between the element.

23

Fig. ‎3.4 Return Loss and Mutual Coupling of Array of two Patches

To steer the beam, phase difference should be inserted between antenna elements. In real

radar system, this task will be handled by phase shifter blocks. But, for simulation purpose,

HFSS has option to insert phase difference between the excitation ports. Figure 3.5 shows

the beam rotation for different values of phase shift. For phase difference of 900 between

exciting ports, the beam rotates by 220 at the H-plane.

24

Fig. ‎3.5 Beam Rotation at H-Plane

As described earlier, patch antennas on high dielectric constant substrates are highly

inefficient radiators due to surface wave losses. This results in a patch antenna with low

gain, efficiency and distorted pattern as well as high level of cross-polarization and mutual

coupling within an array environment. Surface waves are of traveling wave nature and they

can be easily excited inside the silicon substrate. Patch ground plane would be low

impedance medium for surface wave to propagate. So, SWs easily propagates on the

ground plane and cause pattern distortion when they diffract from the edges. One

consequence is that the antenna performance would be dependent on the ground plane size.

Moreover, increasing the size of substrate will distort the pattern more and more. To see the

deteriorating effects of SW, a sample patch antenna is simulated in HFSS for different

ground plane size and also on a larger substrate. Figure 3.6 shows the pattern change at

Eplane for various ground plane size. To add up, the pattern will be the same for Hplane

since SW only propagate in parallel plane to Eplane.

25

(a)

(b)

Fig. ‎3.6 Radiation pattern change at (a) Eplane (b) Hplane

Gain drops 1dB in E & H plane as the ground size enlarges by 3cm in both sides. Pattern

also become more distorted in Eplnae as the ground plane size become larger. The same

problem would be the case in array configuration. The problem caused by surface wave

becomes more sever once the size of substrate grows larger. To see the effect of SW for

26

larger substrate, a comparative simulation is carried out as shown in Figure 3.7. As it can

be seen, pattern is many more distorted for patch antenna with larger substrate.

Fig. ‎3.7 Radiation Pattern (a)Patch with Small Substrate & (b) Patch with Large Substrate

3.1 Solution for Surface Wave Problem

Fabricating microstrip-based planar antennas on Silicon ( ) are strongly preferred

for easy integration with the MMIC/RFIC and RF circuitry. However, as shown in previous

section, patch antennas on high dielectric constant substrates are highly inefficient radiators

due to surface wave losses and have very low gains because of localized EM waves. This

27

results in a patch antenna with low gain, efficiency and distorted pattern as well as high

level of cross-polarization and mutual coupling within an array environment.

In order to tackle the problems caused by surface wave in patch antenna, two approaches

have been pursued to have a patch antenna with optimum performance on high-dielectric

constant substrate. First method uses micromachining technology, while the second makes

use of the concept of electromagnetic band-gap (EBG) structures [5]. In the first approach,

part of the substrate right under the radiating element is removed to establish a low

effective dielectric-constant environment for the antenna. Doing so, power loss due to

surface-wave excitation is reduced and efficiency of energy coupling to space waves

improves. The second approach take advantages of EBG structures: the high-permittivity

substrate is engineered by putting periodic structure to change the propagation

characteristic of a surface wave around antenna operative frequency. Various types of

periodic loading of substrate have been studied [6]. One approach is to drill a periodic

pattern of holes in the substrate or ground plane. Another method is to embed a periodic

pattern of metallic pads inside the substrate; pads are shorted to the ground plane with vias.

In the last one, a type of planar or 2-D loading (no vias are required) were proposed which

is compatible with RFIC integrated circuit fabrication technology (uni-planar

electromagnetic bandgap (Uni-EBG)). Before choosing a method to mitigate surface wave

problem, it will be instructive to study a surface wave nature and its characteristics.

3.1.1 Surface wave:

Surface waves can occur on the interface between two dissimilar materials, such as metal

and free space or dielectric coated conductor. They are bound to the interface, and decay

exponentially into the surrounding materials. At radio frequencies, the fields associated

with these waves can extend thousands of wavelengths into the surrounding space, and they

are often best described as surface currents. Grounded dielectric slab can support surface

waves. These are propagating electromagnetic waves that are bound to the interface

between slab and free space. If the interface surface is smooth and flat, the surface waves

28

will not couple to external plane waves. However, they will radiate vertically if scattered by

bends, discontinuities, or surface texture. Surface waves appear in many situations

involving antennas. On a finite ground plane, surface waves propagate until they reach an

edge or corner, where they can radiate into free space. The result is a kind of multipath

interference which can be seen as ripples in the radiation pattern. Moreover, if multiple

antennas share the same ground plane, surface currents can cause unwanted mutual

coupling [11].

Dielectric coated conductor can support both TM and TE modes [12]. The dominant mode

is TM0 with cutt of frequency of zeros. In other word, it exists in all frequencies. Figure

3.8 shows the dielectric coated conductor together with electric, magnetic field distribution

of dominant mode (TM0) and first mode of patch antenna. As it can seen, the TM0 modes

has the same polarization as that of patch first mode. This shows that TM0 mode can

interfere with first mode of patch if it becomes excited.

Fig. ‎3.8 (a) dielectric coated conductor (b) Electric field(solid line) and magnetic field

distribution of TM0 mode (c) Patch antenna first propagating mode

The cutt of frequency of modes in coated conductor is as below:

( ‎3.1 )

29

As described in previous section, engineered material can be used to filter surface wave and

stop their propagation around frequency of interest.

3.1.2 Application of engineered material to filter surface wave

Metamaterials are engineered structures that exhibit electromagnetic properties not found in

nature. Early in 1968, V. G. Veselago introduced the concept of left-handed (LH) materials,

and described their distinct properties, such as a reversed Snell refraction, inversion of the

Doppler effects, and backward Cherenkov radiation [13]. Metamaterials can be

synthesized by embedding artificially fabricated inclusions in a specified host medium or

on a host surface. Electromagnetic waves interact with the inclusions, inducing electric and

magnetic moments, which in turn affect the macroscopic effective permittivity and

permeability of the bulk composite “medium.” [14].

Meta-materials based structure can control the propagation of wave. This is accomplished

with either a class of Meta-material known as photonic crystals (PC), or another class

known as left-handed materials (LHM); Photonic crystal concept first introduced in optical

region but later it enters the Microwave an MM-wave region where they are called

Electromagnetic band-gap (EBG). Both are a novel class of artificially engineered

structure, and both control and manipulate the propagation of electromagnetic waves

(light). In addition, both EBG and LHM can be designed to have electromagnetic band-gap

at desired frequency. In the EBG structure periodic inclusions inhibit wave propagation due

to destructive interference from scattering from the periodic repetition. An EBG is a result

of a Meta-material that functions in the regime where the period is an appreciable amount

of the wavelength where constructive and destructive interference occur. The band-gap

property of EBG makes them the EM analog of the electronic semi-conductor crystals [15].

Effective permittivity and effective permeability are basic engineering parameters of LHM

metamaterials which is derived from their sub-wavelength structure. Either Negative

permittivity or permeability can inhibit wave propagation. Indeed, the propagation constant

is purely imaginary which result in evanescent mode in structure [16].

30

Microstrip antenna array on high dielectric constant substrate is of special interest due to its

compact size. However, the utilization of high dielectric constant substrate can result in

some drawbacks. One of the severest problems is the inevitable excitation of surface waves,

which can incur strong mutual coupling among elements. The mutual coupling is one of the

major sources of degradation in array performance [6]. Therefore, achieving a high degree

of isolation between elements is an important approach for improving the array

performance. Both EBG and LHM based structure [4-5], [17] can be applied to stop surface

wave propagation around antenna's operative frequency. Hence, decrease the mutual

coupling in the array.

To study MMs, there are two methods. The first choice might be a direct use (or

modification) of a standard free-space method widely used in characterization of

continuous materials. This method involves measurement of the transmission and reflection

coefficients of a slab sample illuminated by a plane wave emanated from a highly directive

antenna. One should use a rather large slab (with typical transversal dimensions of 10

wavelengths) in order to avoid diffraction at the edges. Contrary to the methods with plane-

wave excitation, one may turn to the waveguide methods. The waveguide environment is

well defined and completely closed, the diffraction is not present, and the testing space is

rather small, relaxing the requirements on the sample size. Of course, this environment is

obviously different than free space, and one should be very careful in any interpretation of

the results, particularly in the case of an anisotropic metamaterial [14]. In this project, the

second approach is used.

3.2 Electromagnetic Band-gap Meta-material:

As described earlier, two approaches will be followed to solve the SW problem in patch

antenna. First, part of the substrate right under the radiating element is removed to establish

a low effective dielectric-constant environment for the antenna. Doing so, power loss due to

surface-wave excitation is reduced and efficiency of energy coupling to space waves

improves. Second, EBG structure is used to filter SW: the high-permittivity substrate is

31

engineered by putting periodic structure to change the propagation characteristic of a

surface wave around antenna operative frequency. Various types of periodic loading of

substrate have been studied [6]. One approach is to drill a periodic pattern of holes in the

substrate or ground plane. Another method is to embed a periodic pattern of metallic pads

inside the substrate; pads are shorted to the ground plane with vias. In the last one, a type of

planar or 2-D loading (no vias are required) were proposed which is compatible with RFIC

integrated circuit fabrication technology (uni-planar electromagnetic bandgap (Uni-EBG)).

All in all, there are three type of EBGs reported in literatures which used to control surface

wave propagation. Before going through detailed design procedure for our desired EBG

structure, we will look briefly over different EBG structures and study their characteristics.

3.2.1 Periodic holes:

These types of EBGs are created by drilling a periodic pattern of holes in substrate or

etching a periodic pattern of circles in ground plane. To see the band-gap properties of

EBG, comparative study done for simple mictrostrip transmission line using HFSS

simulator. As shown in Figure 3.9, drilling periodic holes creates a forbidden frequency

band for TEM mode in mictrostrip line. However, drilling cylindrical holes could be

difficult for available foundry process. Moreover, it is possible to drill cubic holes in

substrate but still fabrication cost and difficulty would be limiting factors not to use this

type of EBG structure.

32

Fig. ‎3.9 Microstrip Transmission line with/without EBG

3.2.2 Bumpy Surfaces

Surface waves can be eliminated from a metal surface over a finite frequency band by

applying a periodic texture, such as a lattice of small bumps. As surface waves scatter from

the rows of bumps, the resulting interference prevents them from propagating, producing a

two-dimensional electromagnetic band-gap [11].

The Sievenpiper mushroom structure is an example of such surfaces. It has been widely

studied in the microwave engineering field due to its unique properties. The Sievenpiper

mushroom structure basically consists of a metallic patch connected to ground with a

shorting post. The general model together with HFSS model is shown in Figure 3.10.

33

Fig. ‎3.10 Mushroom EBG structure

It can be used as a high-impedance surface and act like an artificial magnetic conductor.

The Sievenpiper mushroom structure has also been widely used to realize left-handed

Meta-materials (negative refractive index) since it can be designed to support a dominant

quasi-TEM backward wave (i.e. anti-parallel group and phase velocities). It is also evident

from Figure 3.10c that there is weak backward traveling nature associated with this

structure. As depicted in Figure 3.10c the slop of ω-β diagram becomes negative for first

dominant propagation mode. Band-gap property of this type of structure is studied with ω-β

diagram. Method of analysis and the HFSS modeling of this type of EBGs will be describe

in more detail in upcoming sections. To realize this EBG structure metal vias needs to be

inserted inside Silicon substrate after etching is finished and this will make fabrication

process complex and more expensive.

3.2.3 Uni-planar Compact photonic Ban-dgap (UC-PBG):

There is uni-planar compact photonic band-gap (UC-PBG) substrate that has some

advantages over previous structures. Advantageous features of this crystal substrate include

simple low-cost manufacturing (no vias are necessary) and compatibility with standard

monolithic microwave integrated circuits (MMIC’s) fabrication technology. Figure 3.11

shows examples of such planar structure printed on substrate to create forbidden band.

34

Fig. ‎3.11 Tripole Metallodielectric Photonic Band-gap (MPBG) and Uni-EBG structure

This work studies a square type 2-D Uni-EBG structure that is designed specifically to

enhance the performance of microstrip patch antennas at 77GHz. The filtering mechanism

of this structure can be explained with that of LC band-pass filter prototype as shown in

Figure 3.12.

Fig. ‎3.12 Uni-planar Compact EBG Structure Which realizes a 2D Periodic Network of LC

Circuits without Introducing Vias

First, the design procedure of 2-D surface to forbid the propagation of transverse magnetic

(TM) surface waves in a grounded dielectric substrate around antenna's operative frequency

elaborated. It is then demonstrated that a substantial improvement in antenna performance

can be achieved simply by surrounding a microstrip patch antenna with this 2-D Uni-EBG

surface, resulting in a significant increase in both antenna gain and radiation pattern.

35

3.3 Characterization of the 2-D Uni-EBG structure:

An EBG substrate is characterized by a dispersion or ω-β diagram. For a periodic structure

such as the EBG, the field distribution of a surface wave is periodic with a proper phase

delay determined by the wave number k and periodicity p. Moreover, the dispersion curve

for both kx(ω) and ky(ω) is periodic along the k axis with a periodicity of 2π/ px and 2π/ py :

0 ≤ kxn ≤ 2π/px, 0 ≤ kyn ≤ 2π/py , which is known as Brillouin zone [18]. Figure 3.13 shows

the geometry of our 2-D uni-planar EBG surface together with HFSS model for unit cell of

EBG.

Fig. ‎3.13 Uni-planar EBG Surface and HFSS Model for unit cell of Uni-Planar EBG

Surface

Full wave numerical method (FEM of HFSS) has been used to analyze characteristics of

EBG structures with PML and periodic boundary conditions. With the utilization of

periodic boundary conditions (PBCs), only a single cell of the EBG structure needs to be

modeled in full wave simulation [19]. At low frequencies, the impedance is inductive and

structure supports TM surface waves, while in upper bands it supports TE surface waves

36

due to capacitive nature of structure. In between there would be a stop-band in which no

wave propagates. Surface waves with TM nature are responsible for distortion in pattern

since they have same polarization with patch modes and easily harvest energy from patch

mode and block the energy coupling to the space wave. So, our primary goal is to keep

Surface waves with TM nature away from our frequency of interest (77GHz). On the other

hand, surface wave with TE nature have orthogonal polarity with patch mode, so, for the

most part, their impact can be ignored. However, their presence will degrade cross-

polarization. Based on the concepts discussed above, we start to characterize the dispersion

diagram of the Uni-planar EBG structure using the Ansoft HFSS. To see the typical

dispersion diagram, a unit cell of this structure with a= 600um, l=120um, s=278um,

t=80um, g=98um, is simulated on 250um thick Silicon dielectric. Figure 3.14 shows the

dispersion diagram of the surface wave's first three modes for simulated EBG structure.

Fig. ‎3.14 Surface Wave Dispersion Diagram for EBG Structure

37

Increasing the height of the substrate will increase the probability that the surface wave can

be excited. Figure 3.15 shows the dispersion diagram once we double the height of

substrate. Increasing height conceal the semi band-gap region around 70GHz, especially Γ-

X propagation direction. This can be interpreted as increased probability of exciting surface

wave since in thicker dielectric more energy couples to surface wave.

Fig. ‎3.15 Surface Wave Dispersion Diagram for EBG Structure with Doubled Size

An alternative method to model the EBG is by mean of S parameter. To this an ideal

microstrip line can be surrounded by EBG structure and filtering properties can be studied

based on obtained S parameter. Below is the two simulation scenario for characterizing the

EBG structure.

38

Fig. ‎3.16 Two Different Modeling Scheme of EBG to Derive S Parameter

Below is the result for s parameter modeling of EBG structure.

(a)

39

(b)

Fig. ‎3.17 S11 & S21 for (a) Ideal Microstrip Transmission Line (b) Line Surrounded with

EBG Substrate

Figure 3.17 shows not completely but to some extend the filtering property of the EBG

structure. From this simulation result it can be decided that the dispersion or ω-β diagram

characterization yield better result.

3.4 EBG-Patch Antenna design:

Once the proper EBG surface has been designed for surface wave suppression, the design

of an EBG antenna is straightforward. This is done simply by surrounding the antenna with

EBG surface (Figure 3.19). The EBG surface does not interfere with the near field of the

antenna, and it just suppresses the surface waves [15]. However, the presence of the EBG

near the patch drops the resonance frequency which can be removed easily by tuning the

length of the patch. As discussed earlier based on pattern distortion, in patch antenna,

surface wave mostly are present at Eplane. So, it enough for designed EBG only to have

band-gap in this Eplane and filter surface wave with propagation direction in Eplane. Γ-X

40

direction of Brillioum zone is correspondence of patch Eplane and characterizes the surface

wave that exists inside the patch substrate. So, to save computational resources, proposed

EBG will be characterizing in Γ-X plane. Simulation in HFSS is performed as described

earlier for an EBG with band-gap around 77GHz. Figure 3.18 shows the dispersion

diagram of finalized structure with band-gap around our frequency of operation (77GHz).

The simulation is done considering first five modes. Starting from mode one which is of

TM nature then the nature of mode change to TE and continue switching between TM and

TE. The nature of first mode can be attributed to the inductive property of the structure at

low frequency. Remembering the LC band-pass filter model described earlier, one can

discern why this structure is inductive at low frequency. Moreover, the mode propagation

diagram is trimmed by light line since for the propagation constant smaller than that of line

light the mode will be evanescent. Desired band-region is shown in Figure 3.18 that is

between 72-77GHz.

Fig. ‎3.18 Dispersion Diagram for EBG with Band-gap around 77GHz

Band-gap Region

41

The next step is to surround the patch antenna with proposed EBG Substrate and check for

it performance. Figure 3.19 shows patch antenna with EBG around together with antenna's

S11.

Fig. ‎3.19 Designed Antenna with its S11

42

For comparison purposes, another patch antenna designed on a same substrate without EBG

surface. Figure 3.20 compares the radiation pattern and gain of these antennas, both in the

E and Hplane.

Fig. ‎3.20 Radiation pattern for two type of antenna

Designed antennas are submitted for fabrication and next chapter will provide the

measurement result.

3.5 Antenna integration with active circuitry

As described earlier, radar system RF front end consist of array of antenna with LNAs and

phase shifter. The output of the antenna will be amplified by LNA circuit. For the antenna

pattern steering, a 2x1 array is designed and each antenna line has a very low loss MEMS

based phase shifter. Using these MEMS phase shifters, the antenna main beam will be

steerable. Also, one can appreciate the very low loss phase shifters, since at these

frequencies; signal levels are already too low. At these frequencies, MEMS based phase

shifter is preferred due to their low loss. Using IHP MEMS technology, it is already shown

that low loss around 0.5 dB can be obtained. After the antenna, LNA and the phase shifter,

43

a corporate type power combiner will be used to combine the signals coming from the two

antennas. All the antennas, phase shifters, combiner and the LNA will be on the same

single chip.

An array of two patch antennas integrated with LNA and phase shifter is submitted to be

fabricated by IHP.

Fig. ‎3.21 Layout of Phased Array RF Front-End

Patch Antenna

Patch Antenna

LNA

LNA

MEMS Phase Shifter

44

4. Antenna Fabrication & Measurement Result

Millimeter wave technology brings about many promising feature such as possibility of

high speed communication, less interference, etc. However, conducting practical

experiments at this frequency band could be challenging and will require high-tech devices

and skilled experimenters. First challenge is the high losses which make it necessary to

consider transmitted power level, proper signal amplification and component spacing. For

antenna measurement, anechoic chamber can be used up for frequency up to 50GHz.

However, for 75-110GHz, a setup similar to that of anechoic chamber needs to be

implemented outside the chamber. Outdoor setup occupies less space and also makes it

possible to better connect the antenna to measuring probes. Our antenna measurement setup

at Sabanci University seeks to fulfill these necessities and make reliable measurement

possible. To do so, our measurement setup follows all the standard procedure in antenna

measurement. Figure 4.1 shows schematic of outdoor antenna setup block that consists of

Network analyzer, standard horn antenna, extender, GSG probe and rotatable arms.

45

Moreover, waveguide to coaxial adaptor is used in different part for transition purpose.

Measurement follow the same rule applied in anechoic chamber.

Fig. ‎4.1 Schematic of Millimeter Wave Antenna Setup

GSG probe is used to connect the antenna under test (AUT) to signal source. There are

knobs places on GSG probe for fine adjustment in different direction. GSG probe is

connected to Network analyzer (NA), with extender block in between. Extender serves the

purpose of signal frequency up conversion to convert 300MHz-50GHz signal from NA to

75-110GHz necessary for W-band measurement. To measure the field radiated by AUT, a

standard horn with known gain is used. This horn is installed on collecting arm capable of

rotating 1800 at elevation plane. Care should be taken to place horn antenna in far field

region of AUT not too far to lose signal level. Another type of GSG probe is available with

900 rotations with previous one. This probe make it possible to put AUT in 90

0 rotated with

respect to previous case and switch from antenna's E to H plane and vice versa. Besides, the

standard horn can be rotated in 900 to its original position and make it possible to determine

46

the co- and cross-polarization at E and H plane. This setup enables reflection coefficient,

gain and far-field radiation pattern measurement. Our antenna under the test (AUT) is

dipole antenna and simple inset-fed patch antenna surrounded by EBG structure. A sample

of the antennas has been realized by MMIC technology available at IHP, Germany. Figure

4.2 shows an implemented setup picture at SUNUM.

Fig. ‎4.2 Implemented Millimeter Wave Antenna Setup

4.1 Simulated and measured result:

At the time of writing this thesis, we were waiting for our patch antenna from IHP to be

shipped to us. But, to see the functionality of the implemented setup, a sample of dipole

antenna and patch antenna from previous works has been measured by setup described

47

earlier. Figure 4.3 shows the measured and simulated result for dipole antenna printed on

etched silicon substrate. As it can be seen, there is a frequency shift in response of the

measured result. This change can be attributed to uncertainty about exact value of Silicon

dielectric constant. There is no reliable reference presenting exact value of silicon

dielectric constant at W-band. To have correct model of substrate at simulation, dielectric

constant needs to be determined. So, the next section of this thesis will be about finding

value of silicon dielectric constant based of available methods.

Fig. ‎4.3 Simulated and Measured S11 for dipole antenna

Figure 4.4 shows simulated and measured result for a sample patch antenna from previous

work.

48

(a) Simulated patch antenna with its reflection coefficient

(b) Fabricated patch antenna with its reflection coefficient

Fig. ‎4.4 Measured and simulated result for sample patch antenna

4.2 Dielectric constant measurement:

There is uncertainty about the dielectric constant of substrates such as silicon at W-band

frequencies. Silicon dielectric uncertainty effect becomes very important for the design and

30 40 50 60 70 80 90 100-10

-9

-8

-7

-6

-5

-4

-3

-2

-1

Freq(GHz)

Re

turn

Lo

ss

S11 Simulated Patch antenna

75 80 85 90 95 100 105 110-20

-17.5

-15

-12.5

-10

-7.5

-5

Freq(GHz)

Retu

rn

Loss

Sample Patch 1

Sample Patch 2

49

simulations of RF circuits and antennas on silicon substrates. Designing an antenna on a

substrate with inaccurate value of dielectric constant will change antenna performance and

shift the resonance frequency in practice which causes large discrepancies between

simulated and measured results. Simple and applicable methods for measuring the

permittivity of the substrate are always in great interest of the microwave circuit and

antenna designers.

There are various methods for measuring dielectric constant including dielectric

waveguide, cavity resonator, open resonator and free space method which is more suitable

for wideband measurement [7]. In free space method, the dielectric constant is measured

based on either the transmission method or metal backed method [8-9]. Metal backed

method is better for very thin dielectric samples. However, some disadvantages such as

edge diffraction and multiple reflections are associated with the free space method. Edge

diffraction can be minimized by using spot focus antenna which produce Gaussian beam by

mean of lens and also taking the sample size large enough, while time domain filtering and

LRL (line, reflect, line) calibration technique can be used for multiple reflection problem.

Using amplitude and phase information it is possible to derive both real and imaginary

parts of dielectric constant. However, it is possible to derive only the real part just using the

phase information of either transmitted or reflected signal. In this work, two approaches are

applied to bring out the real part of silicon dielectric constant based on transmission and

reflection type measurement method. Note that no calibration and time domain filtering is

utilized in the measurements. The transmitter and receiver antennas which are used in the

measurement set-up have wide beamwidth far away from Gaussian beam. Only the phase

information of the transmitted signal is used to determine the real part of the permittivity.

4.2.1 Theory of dielectric constant derivation

Travel time of EM wave is different for different Medias and this time difference can be

interpreted as phase difference in frequency domain. A slab of silicon wafer with dielectric

50

constant of ε1 is placed in the far fields of the two horn antennas as shown in Figure 4.4.

Thickness of the DUT is supposed to be d1, while d0+d1 is the total distance between the

two antennas. Impedance variation in the direction of wave propagation will cause multi-

reflections inside the silicon sheet and infinite series of rays with different phase difference

will be picked up by the receiver antenna (Ant. 2 in Fig. 1). S21 between two antennas can

be calculated for the configuration of the Figure 4.4 which is confined to the air (ε0) at

both sides of the DUT.

Fig. ‎4.5 Schematic of the Measurement Setup

Analytical relation for S21 is obtained as follow,

0 0 0 01 1 1 1

0 0 1 1 1 1 1 1

0 0 1 1 1 1

3

2 1 2 1 1 2 2 1 1 2 1 2 1 2

( ) 2 42 4

2 1 1 2

( ) 22

2 1 1 2

(1 ...)

(1 ) , 1

jk d jk djk d j k d

j k d k d j k d j k d

j k d k d j k d

S e T e T e T e T

e T T e e

e T T e

( ‎4.1)

where T21 and T12 are transmission coefficients from air to dielectric and vice versa,

respectively.

Г12 is reflection coefficient from dielectric to air and is defined as:

1 2

1

1

r

r

( ‎4.2)

51

and

0, 0 ,1

i ik i

( ‎4.3)

which is wave-number in each of the media.

Accurate measured phase between transceiver antennas will be

2 10 0 1 1

1 2

1 1 1 1

( )

ta n (s in ( 2 ) ( c o s ( 2 )) )

Sk d k d

k d k d

( ‎4.4)

which is deduced from relation (4.1). We can use phase of the measured S21 and (4.4) with

determined d0+d1 and d1 to calculate the permittivity of silicon at the measured frequency.

In this case, accurate values of measured phase, transceiver antennas spacing and dielectric

thickness is needed to achieve accuracy in the calculated .

In the proposed method, phase of S21 is measured in two cases; 1) Silicon sheet is placed

between the antennas. 2) Silicon is absent between the antennas.

Phase difference between two cases can be calculated by subtracting relation in (4.4) when

the silicon is between the antennas and silicon is replaced by air (by setting εr =1 or Г=0 in

(4.4)).

2 1 2 11 0 1

.

1 2

1 1 1 1

( )

tan (s in ( 2 ) ( co s ( 2 )) )

S Sa ir su b

k k d

k d k d

( ‎4.5)

The calibrated formula is independent of the spacing between the antennas. If multiple

reflections are neglected, one can obtain a closed form expression for the solution which is

given by

52

1 0 1 1( ) 2 ( 1)

rk k d fd c

( ‎4.6)

by replacing Г=0 in (4.5). Although evaluated εr in (4.6) cannot achieve a good

approximation due to neglect multiple reflections, but, (4.6) can easily be solved for

relative dielectric constant as given in (4.7).

2

1(1 (2 ) )

rc fd

( ‎4.7)

To suppress the effect unwanted reflection absorber material can be used to make a

reflection free environment for measurement. Figure 4.5 shows the implemented setup.

Two standard horn antenna used as transmit and receive antenna. Absorber material is

placed in different side to avoid undesired reflections from floor and surrounding objects.

These reflections can cause error in measured result. Another method to omit unwanted

reflections is to use time domain gating which is possible with PNA network used in our

measurement. However, only absorber material is used to get rid of reflections. The

measurement is done using the setup implemented as in Figure 4.5. The phase difference

between Silicon slab with 500um and air is shown in Figure 4.6 and corresponding

dielectric constant is depicted in Figure 4.7.

53

Fig. ‎4.6 Implemented Dielectric Setup

54

Fig. ‎4.7 Measured Phase Difference versus Frequency

55

Fig. ‎4.8 Calculated dielectric constant

Calculated dielectric constant for Silicon can be used for later simulation modeling at full

wave simulators. This will allow avoiding frequency shift in response of simulated and

measured result.

56

CONCLUSION AND FUTURE WORK

In this thesis a design of reliable antenna system for automotive radar system at 77GHz is

studied. Various type of antenna including dipole antenna and patch type antenna can be

used for this purpose. The choice of the antenna is done based on the radar system range

(gain requirements), ease of fabrication, possibility of integration with other circuitry and

cost. At 77GHz since the size of antenna is small, it is possible to think about on chip

antenna design which can integrate the antenna and rest of RF circuitry and make radar

compact and cost effective. For this purpose, a design of different antenna like patch,

dipole, slot type antenna on Silicon substrate is studied. Patch antenna is selected for final

assembly based on the criteria mentioned for 77GHz radar system like gain and fabrication

consideration. However, the original antenna on silicon substrate suffers from low gain due

to Silicon loss and gain and pattern distortion due to excitation of surface wave. Two

approaches is taken to overhaul the problems. First, available etching technique is used to

57

remove the substrate under the radiating antenna to improve the radiation properties.

Second, Electromagnetic band-gap structure is used to filter the unwanted surface wave

inside the Silicon substrate. Moreover, further gain improvement is done by array of

proposed patch antenna. This also gives the radar system the beam steering option. All the

simulations are carried out by FEM based simulator, HFSS. The accuracy of the simulated

result is verified with antenna measurement setups implemented at SUNUM laboratory.

Antenna setup enables us to conduct S parameter, impedance and radiation pattern

measurement. Conducting measurement at this frequency range is a little bit of challenge

and requires high tech devices and good practice. A sample of single antenna and antenna

in array configuration are realized at IHP Company. Measured results for reflection

coefficient of sample dipole antenna from previous works reveal that there is frequency

shift between simulated and measured result. This shift can be attributed to the uncertainty

about dielectric constant of the Silicon at W-band frequency. In simulation, dielectric

constant is assumed to be 12 (low frequency model) for Silicon. However, the difference

between simulated and measured result reveals stringent need for knowledge about exact

value of Silicon dielectric constant at W-band. So, another chapter is devoted to study the

methods of determining the exact value of the Silicon dielectric constant at W band

frequency range. Among various techniques, free space method is chosen with two

standard horn antennas as transmitter and receiver. Propose free space method is based on

phase information of transmitted signal for different case; the silicon slab between and air.

From the phase difference between these two measurements, the dielectric constant of

silicon is derived. Advantage of free space is its wideband operation and simple

implementation. A dielectric measurement setup is prepared to do the experiment.

Measured result seems to properly model for silicon. This result is used to accurately model

Silicon substrate for next simulations.

Throughout this thesis, we tried to design reliable antenna system for 77GHz radar system

and the focus was on designing on chip antenna. However, antenna design on substrate like

Silicon has its own drawbacks. Proposed solutions can mitigate the problems not

completely but to some extent. Moreover, the cost of realizing overall chip at foundry is

58

high for moderate size chip and the price keeps increasing exponentially with the size of

chip. Future research study can be focus on designing on chip antenna by replacing the

silicon with another low loss, low dielectric constant substrate like FR4. This is possible by

using wire-bonding techniques to still keep the single chip solution valid for proposed

circuitry. For example, the active circuitry can be realized by foundry process and antenna

can be fabricated with low cost printed board technologies. For final assembly, antenna and

active part can be connected by wire-bonding techniques.

59

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